Power Electronics Technology

TECHNOLOGY 

® 

THE ENGINEER’S SOURCE FOR POWER AND ENERGY EFFICIENCY DESIGN INFORMATION 

Metal in 

the Board 

Boosts PCB 

Power Density 

p. 8 

www.powerelectronics.com 

JUNE 2013 

Vol. 39, No. 6 

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EDITOR’Sviewpoint 

Three-Engineer Team Wins $1M for 

DARPA’s Ground Vehicle Challenge 

THE DEFENSE Advanced Research Projects 

Agency (DARPA) awarded a $1 million prize 

to “Ground Systems”, a three-person team with 

members in Ohio, Texas and California, as the 

winner of the Fast Adaptable Next-Generation 

Ground Vehicle (FANG) Mobility/Drivetrain Challenge. 

Team Ground Systems’ final design submission received the 

highest score when measured against the established requirements 

for system performance and manufacturability. 

“I’m very pleased with the quality of the submissions we 

received during the challenge, and we have learned a great 

deal throughout the process,” said Army Lt. Col. Nathan 

Wiedenman, DARPA program manager. “The first FANG 

Challenge has been a great experiment, and the submission 

of many viable, innovative designs has validated the 

Adaptive Vehicle Make (AVM) design tools and provided 

invaluable feedback to continue their development.” 

Wiedenman noted that several different types of teams 

were able to use various aspects of the tools to create viable 

designs in the course of the challenge. The winning team, 

for example, was geographically separated, but was able 

to use the collaboration tools to create the winning design. 

Another finalist team was comprised of people who met 

through VehicleFORGE, the online collaboration platform 

used by competitors to manage and submit their designs. 

Still another top design was submitted by a one-person team. 

In many cases, a traditional design process would likely have 

excluded these teams from contributing their ideas. 

Since the beginning of the first FANG Challenge on 

January 14, 2013, more than 1,000 participants within 

more than 200 teams used the META design tools and 

the VehicleFORGE collaboration platform developed by 

Vanderbilt University in Nashville, Tenn., to design and 

simulate the performance of thousands of potential mobility 

and drivetrain subsystems. The goal of the FANG program 

is to test the specially developed META design tools, model 

libraries and the VehicleFORGE platform, which were created 

to significantly compress the design-to-production time 

of a complex defense system. 

Now that the design challenge has concluded, the winning 

FANG design will be built by the DARPA iFAB program 

team. iFAB, or Instant Foundry Adaptive through Bits, is led 

by the Applied Research Laboratory at Penn State University 

and will validate the manufacturability feedback, foundry 

configuration, and instruction generation tools as part of the 

build process. Ultimately, the as-built design will be subjected 

to test and evaluation under the leadership of the FANG performer, 

Ricardo Inc. of Van Buren Township, Mich. 

Begun in 2010 as part of DARPA’s advanced manufacturing 

initiative, AVM is a portfolio of programs focused 

on the reduction of complex military system development 

timelines by a factor of five or more. The technical approach 

encompasses multiple efforts addressing all aspects of the 

manufacturing process, from requirements representation, 

through design, to final physical build of a full-scale complex 

defense system. 

The ultimate goal of the META program is to dramatically 

improve the existing systems engineering, integration, 

and testing process for defense systems. META is not predicated 

on one particular alternative approach, metric, technique 

or tool. Broadly speaking, however, it aims to develop 

model-based design methods for cyber-physical systems far 

more complex and heterogeneous than those to which such 

methods are applied today; to combine these methods with 

a rigorous deployment of hierarchical abstractions throughout 

the system architecture; to optimize system design with 

respect to an observable, quantitative measure of complexity 

for the entire cyber-physical systems; and to apply probabilistic 

formal methods to the system verification problem, 

thereby dramatically reducing the need for expensive realworld 

testing and design iteration. 

VehicleFORGE aims to significantly expand open source 

collaborative development for defense systems by employing 

a general representation language — being developed under 

the META program. 

SAM DAVIS, Editor-in-Chief 

2 Power Electronics Technology | June 2013 www.powerelectronics.com

EDITORIAL 

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www.powerelectronics.com June 2013 | Power Electronics Technology 3

CoverStory 

p.7 

Metal in the Board: Meeting 

The Challenge of Digital Power 

By Nick Pearne and Bill Burr, BPA Consulting. 

New generations of miniaturized power 

packages require thermal and electrical 

pathways into the board. Described here are 

these developments and a design example 

for an innovative printed circuit approach for 

a light electric vehicle powerpack. 

COVER CREDIT: Fraunhofer IIS Battery Management System 

with active cell control andsemiconductor safety shutdown 

(courtesy of Häusermann GmbH) 

FOR DESIGNERS AND SYSTEMS ENGINEERS 


JUNE 2013 • Vol. 39, No. 6 

DESIGN FEATURES 

Stability Criteria Of 

A Control System 

13 

Excerpts from the book “Designing 

Control Loops for Linear and Switching 

Power Supplie”, focusing on what engineers 

really need to know to compensate 

or stabilize a given control system. 

eGaN® FET-Silicon Power 

19 Shoot-Out: Small Signal 

RF Performance 

This article of the eGaN® FET-Silicon 

Shoot-Out series examines RF performance 

of the EPC2012 eGaN FET. 

Back to Basics: Voltage 

24 Regulator ICs, Part 1 

A primer on the two major types of regulator 

ICs, linear and switch-mode. 

Intelligent Energy 

29 Optimization System 

Slashes Electric Bills 

The SP1000 series of hybrid energy 

management and correction systems 

guarantees and quantifi es kilowatt-hour 

savings by reducing power delivery 

losses. 

SMA 

Connector – 

Gate 

Connection 

(AC & DC) 

Thermal Sensor 

Connection 


D.U.T. (e.g. EPC2012) 50 Ω Micro-Strip 

CURRENT TRENDS 

PET INNOVATIONS 

Gas Gauge IC Monitors 

31 Lead-Acid Battery State-Of- 

Health, State-Of-Charge 

An accurate, simple-to-use gas gauge 

IC monitors lead-acid batteries used in 

mobile and stationary applications. 

DEPARTMENTS 

2 EDITOR’S VIEWPOINT 

6 INDUSTRY HIGHLIGHTS 

35 NEW PRODUCTS 

38 PATENTS 

40 PRODUCT MARKETPLACE 

40 CLASSIFIED ADS 

40 ADVERTISER INDEX 

SMA 

Connector – 

Drain 

Connection 

(AC & DC) 

4 Power Electronics Technology | June 2013 www.powerelectronics.com 

Hea 

Mounti

CoverStory 

p.7 

Metal in the Board: Meeting 

The Challenge of Digital Power 

By Nick Pearne and Bill Burr, BPA Consulting. 

New generations of miniaturized power 

packages require thermal and electrical 

pathways into the board. Described here are 

these developments and a design example 

for an innovative printed circuit approach for 

a light electric vehicle powerpack. 

COVER CREDIT: Fraunhofer IIS Battery Management System 

with active cell control andsemiconductor safety shutdown 

(courtesy of Häusermann GmbH) 

FOR DESIGNERS AND SYSTEMS ENGINEERS 


JUNE 2013 • Vol. 39, No. 6 

DESIGN FEATURES 



13 

Excerpts from the book “Designing 

Control Loops for Linear and Switching 

Power Supplie”, focusing on what engineers 

really need to know to compensate 

or stabilize a given control system. 

eGaN® FET-Silicon Power 

19 Shoot-Out: Small Signal 

RF Performance 

This article of the eGaN® FET-Silicon 

Shoot-Out series examines RF performance 

of the EPC2012 eGaN FET. 

Back to Basics: Voltage 

24 Regulator ICs, Part 1 

A primer on the two major types of regulator 

ICs, linear and switch-mode. 

Intelligent Energy 

29 Optimization System 

Slashes Electric Bills 

The SP1000 series of hybrid energy 

management and correction systems 

guarantees and quantifi es kilowatt-hour 

savings by reducing power delivery 

losses. 

SMA 

Connector – 

Gate 

Connection 

(AC & DC) 


Connection 



CURRENT TRENDS 

PET INNOVATIONS 

Gas Gauge IC Monitors 

31 Lead-Acid Battery State-Of- 

Health, State-Of-Charge 

An accurate, simple-to-use gas gauge 

IC monitors lead-acid batteries used in 

mobile and stationary applications. 

DEPARTMENTS 

2 EDITOR’S VIEWPOINT 

6 INDUSTRY HIGHLIGHTS 

35 NEW PRODUCTS 

38 PATENTS 

40 PRODUCT MARKETPLACE 

40 CLASSIFIED ADS 

40 ADVERTISER INDEX 

SMA 

Connector – 

Drain 

Connection 

(AC & DC) 


Hea 

Mounti

INDUSTRY 

APEC 2014 Call for Papers Now Open 

APEC 2014 CONTINUES the long-standing tradition 

of addressing issues of immediate and longterm 

interest to the practicing power electronic engineer. 

For more information visit our website or view 

the full Call for Papers. Topics Papers 

of value to the practicing engineer are 

solicited in the following topic areas: 

• AC-DC and DC-DC Converters 

• Power Electronics for Utility Interface 

• Motor Drives and Inverters 

• Devices and Components 

• System Integration 

• Modeling, Simulation, and Control 

• Manufacturing and Business Issues 

ROBOTIC INSECTS MAKE FIRST CONTROLLED FLIGHT 

THE DEMONSTRATION OF THE FIRST 

CONTROLLED FLIGHT of an insectsized 

robot is the culmination of more 

than a decade’s work, led by researchers 

at the Harvard School of Engineering and 

Applied Sciences (SEAS) and the Wyss 

Institute for Biologically Inspired Engineering 

at Harvard. 

“This is what I have been trying to do 

for literally the last 12 years,” says Robert J. 

Wood, Charles River Professor of Engineering 

and Applied Sciences at SEAS, Wyss Core 

Faculty Member, and principal investigator of 

the National Science Foundation-supported 

RoboBee project. “It’s really only because of 

this lab’s recent breakthroughs in manufacturing, 

materials, and design that we have 

even been able to try this. And it just worked, 

spectacularly well.” 

Inspired by the biology of a fly, with submillimeter-scale 

anatomy and two wafer-thin 

wings that flap almost invisibly, 120 times 

per second, the tiny device not only represents 

the absolute cutting edge of micromanufacturing 

and control systems; it is an 

aspiration that has impelled innovation in 

these fields by dozens of researchers across 

Harvard for years. 

“We had to develop solutions from 

scratch, for everything,” explains Wood. “We 

would get one component working, but when 

we moved onto the next, five new problems 

would arise. It was a moving target.” 

Flight muscles, for instance, don’t come 

prepackaged for robots the size of a fingertip. 

“Large robots can run on electromagnetic 

motors, but at this small scale you have to 

come up with an alternative, and there wasn’t 

one,” says co-lead author Kevin Y. Ma, a 

graduate student at SEAS. 

The tiny robot flaps its wings with piezoelectric 

actuators-strips of ceramic that 

expand and contract when an electric field 

is applied. Thin hinges of plastic embedded 

within the carbon fiber body frame serve as 

joints, and a delicately balanced control system 

commands the rotational motions in the 

flapping-wing robot, with each wing controlled 

independently in real-time. 

At tiny scales, small changes in airflow 

can have an outsized effect on flight dynamics, 

and the control system has to react that 

much faster to remain stable. 

The robotic insects also take advantage 

of an ingenious pop-up manufacturing technique 

that was developed by Wood’s team in 

Power Electronics Applications Important Dates 

• Deadline for digest submission is July 8, 2013 

• Notification that a submission was accepted or declined 

will be emailed no later than October 7, 2013. 

• Final submission and registration will be 

accepted no later than November 18, 2013. 

• At APEC 2014, the technical sessions 

will be given on March 18 - 20, 2014 

Upload your submission online at http://www.epapers. 

org/apec2014 no later than Monday, July 8, 2013. 

• Call For Reviewers: Please sign-up to become a technical 

paper reviewer by July 12th. Accepted reviewers will be 

notified by August 5th. All reviews must be completed by 

September 2nd. 

2011. Sheets of various laser-cut materials 

are layered and sandwiched together into 

a thin, flat plate that folds up like a child’s 

pop-up book into the complete electromechanical 

structure. 

The quick, step-by-step process replaces 

what used to be a painstaking manual art and 

allows Wood’s team to use more robust materials 

in new combinations, while improving 

the overall precision of each device. 

“We can now very rapidly build reliable 

prototypes, which allows us to be more 

aggressive in how we test them,” says Ma, 

adding that the team has gone through 20 

prototypes in just the past six months. 

Applications of the RoboBee project could 

include distributed environmental monitoring, 

search-and-rescue operations, or assistance 

with crop pollination, but the materials, fabrication 

techniques, and components that 

emerge along the way might prove to be even 

more significant. For example, the pop-up 

manufacturing process could enable a new 

class of complex medical devices. Harvard’s 

Office of Technology Development, in collaboration 

with Harvard SEAS and the Wyss 

Institute, is already commercializing some of 

the underlying technologies. 


DESIGNfeature 

NICK PEARNE AND BILL BURR, BPA Consulting 

METAL IN THE BOARD: 

MEETING THE CHALLENGE OF 

DIGITAL POWER 

New generations of miniaturized power packages require thermal and electrical pathways 

into the board. Described here are these developments and a design example for a new and 

innovative printed circuit approach for a 150 – 300 A light electric vehicle powerpack. 

New families of power semiconductors 

require new approaches in board design 

for thermal and power management. A 

number of board level solutions have 

emerged, divided into 12 major types 

segregated by: 

• Thermal management technology- heat pipes, inlays, dissipation 

planes, etc. 

• Current management- copper planes, embedded bus 

bars, discrete wiring or strips 

• Board layup- number of layers, use of internal/external 

dissipator planes 

The basic building blocks for “Metal in the Board” or 

“MiB” consist of: 

• Metal based laminate 

• Thick copper layers 

• Thermal vias 

• Thermal risers 

• Metal inlays 

• Encapsulated bus bars 

• Bonded and/or embedded wires and strips 

These building blocks offer a wide range of thermal 

management capabilities, and have been characterized in 

terms of the power densities they are capable of managing 

with less than 10 °C temperature rise through the board 

thermal path. Solutions typically range from 0.25 W/cm 2 

up through 35 W/cm 2 - through the board. This range of 

capabilities is proving to be critical as power conversion, 

management, and control becomes digital and advances in 

power semiconductor technology combined with the cost 

advantages of automated assembly drive development of 

new and smaller surface mount packaging. 

HIGH POWER 

One of these is the “isometric” family of packages, developed 

by International Rectifier and marketed as the 

“DirectFET®.” A similar package is offered under license 

to IR by Infineon- the “CanPAK”. These devices are 

“isometric” because they provide a balanced thermal pathway 

both into the board and, if necessary, out through the 

top of the package. This represents a significant advantage 

over other SMT power semiconductor packages such as 

the TO-252 “D2PAK” or TO-263 types, which require 

“gullwing” heat sinks if the board design is not capable of 

supplying a sufficiently low thermal resistance pathway 

(Fig. 1). 

While the isometric types offer greater flexibility in 

management of the thermal path, they are also capable of 

currents well in excess of the 10 – 15 A typically considered 

the top end for conventional printed circuit boards. 

The power applications typically serviced by MOSFETs 

or IGBTs in isometric packaging involve currents from 50 

to several hundred amperes, and managing these current 

levels involves a different and unique set of board design 

criteria. 

AMPACITY 

Current flow through a conductor causes resistive power 

losses (I 2 R) in the form of heat, and at high current lev- 


THERMALmanagement 

els the temperature increase 

becomes a factor in determining 

ampacity because the 

resistivity of the conductor 

changes with temperature. 

The relationship is linear, i.e., 

resistivity increases proportional to the change in temperature 

at a rate determined by the temperature coefficient of 

the conductor: 

R T = R T0 × [(1 + (T - T 0 )] (1) 

Where: 

T = Temperature at which resistivity is measured 

T 0 = Reference temperature (ambient) 

= Linear temperature coefficient (copper = 0.004) 

R T = Resistivity at measurement temperature 

R T0 = Resistivity at reference temperature 

For a copper conductor, every 25°C increase in temperature 

means a drop of about 5% in maximum ampacity 

due to an increase in conductor resistivity, RT . Since 

this presents the probability of further power dissipation 

and temperature rise, MiB design practice must consider 

effective methods not only to control and reduce conductor 

resistivity, but also to provide low thermal resistance 

pathways for heat dissipation. 

In a printed circuit board, ampacity depends on a number 

of different factors: 

• Conductive + convective capability provided by spreading 

layers, ground layers, stack-up 

• Ratio of track width to thickness 

• Ambient temperature 

• Adjacent high current tracks 

• AC or DC current 

• Presence and frequency of partial crosssection 

shrinkage 

• Presence, number, and conductive 

cross-section of plated through holes in 

series with the conductor. 

Therefore effective design needs to 

consider more variables than are normally 

addressed by the IPC-2152 current vs. 

temperature charts. 

MIB 

“Metal in the Board” or “MiB” includes 

a number of approaches where MiB 

building blocks are combined to provide 

effective high-current solutions. 

The “DWPCB” (discrete wire PCB) type 

Fig. 1. Gullwing heat sink on TO-252 

power device. 

External Only 

Low heat spreading 


Wire Bonder 

Bonded Elements 

Etched Conductor 

Pattern 

Substrate Cross-Section 

Fig. 2. “Discrete Wire” process- bonding high current elements to inner layer. 

is one of the most versatile. One of the commercially 

available versions of DWPCB is HSMtec, developed by 

the Austrian PCB manufacturer Häusermann GmbH. 

HSMtec uses 0.5 mm diameter copper wire and rectangular 

sectioned 0.5 mm thick copper strips (“profiles”) 

to provide discrete low resistance electrical and thermal 

pathways in the board as shown in Fig. 2. 

There are a number of advantages to this solution compared 

to conventional thick copper or metal core boards: 

• Conventional PCB processes ensure consistently high 

reliability 

• Enhanced thermal and current pathways only where 

needed 

• Cost of MiB limited to those nets needing MiB 

• Wiring densities up to and including HDI enable logic 

and power integration 

• FR-4 materials reduce CTE mismatch common to aluminum 

based substrates 

• Board may be folded during assembly, providing photometric 

solutions for LED luminaires and eliminating 

daughter boards/connectors 

The profiles and wires that make up the MiB compo- 

Internal + External 

Heat spreading 


Fig. 3. Effect on heat spreading of thermal dissipation plane (source: Häusermann GmbH). 



els the temperature increase 

becomes a factor in determining 

ampacity because the 

resistivity of the conductor 

changes with temperature. 

The relationship is linear, i.e., 

resistivity increases proportional to the change in temperature 

at a rate determined by the temperature coefficient of 

the conductor: 

R T = R T0 × [(1 + (T - T 0 )] (1) 

Where: 

T = Temperature at which resistivity is measured 

T 0 = Reference temperature (ambient) 

= Linear temperature coefficient (copper = 0.004) 

R T = Resistivity at measurement temperature 

R T0 = Resistivity at reference temperature 

For a copper conductor, every 25°C increase in temperature 

means a drop of about 5% in maximum ampacity 

due to an increase in conductor resistivity, RT . Since 

this presents the probability of further power dissipation 

and temperature rise, MiB design practice must consider 

effective methods not only to control and reduce conductor 

resistivity, but also to provide low thermal resistance 

pathways for heat dissipation. 

In a printed circuit board, ampacity depends on a number 

of different factors: 

• Conductive + convective capability provided by spreading 

layers, ground layers, stack-up 

• Ratio of track width to thickness 

• Ambient temperature 

• Adjacent high current tracks 

• AC or DC current 

• Presence and frequency of partial crosssection 

shrinkage 

• Presence, number, and conductive 

cross-section of plated through holes in 

series with the conductor. 

Therefore effective design needs to 

consider more variables than are normally 

addressed by the IPC-2152 current vs. 

temperature charts. 

MIB 

“Metal in the Board” or “MiB” includes 

a number of approaches where MiB 

building blocks are combined to provide 

effective high-current solutions. 

The “DWPCB” (discrete wire PCB) type 

Fig. 1. Gullwing heat sink on TO-252 

power device. 

External Only 



Wire Bonder 

Bonded Elements 

Etched Conductor 

Pattern 

Substrate Cross-Section 

Fig. 2. “Discrete Wire” process- bonding high current elements to inner layer. 

is one of the most versatile. One of the commercially 

available versions of DWPCB is HSMtec, developed by 

the Austrian PCB manufacturer Häusermann GmbH. 

HSMtec uses 0.5 mm diameter copper wire and rectangular 

sectioned 0.5 mm thick copper strips (“profiles”) 

to provide discrete low resistance electrical and thermal 

pathways in the board as shown in Fig. 2. 

There are a number of advantages to this solution compared 

to conventional thick copper or metal core boards: 

• Conventional PCB processes ensure consistently high 

reliability 

• Enhanced thermal and current pathways only where 

needed 

• Cost of MiB limited to those nets needing MiB 

• Wiring densities up to and including HDI enable logic 

and power integration 

• FR-4 materials reduce CTE mismatch common to aluminum 

based substrates 

• Board may be folded during assembly, providing photometric 

solutions for LED luminaires and eliminating 

daughter boards/connectors 

The profiles and wires that make up the MiB compo- 

Internal + External 



Fig. 3. Effect on heat spreading of thermal dissipation plane (source: Häusermann GmbH). 



Layer Stack 

Δ T (°C) 

20 

40 

60 

80 

100 

120 

Round wire 

0.5 mm 

8.3 26.6 

11.8 

14.4 

16.7 

18.6 

20.4 

Cu – Profile 

2 mm 

37.7 

46.1 

53.3 

59.6 

65.3 

FR4 160 mm x 100 mm x 1.6 mm 

Cu - Profile 

4 mm 

44.0 

62.3 

76.3 

88.1 

98.5 

107.9 

Cu -Profile 

8 mm 

nents in a DWPCB board are bonded to tracks etched on 

inner layer cores, basically forming a sandwich consisting 

of the etched track and the bonded elements. This patented 

process ensures the uniform contact between the track 

and the wire/profile, which is essential for homogenous 

heat spreading and uniform conductor cross-sectional area. 

It also simplifies the layout task and/or conversion from 

conventional designs, as placement of the high current 

MiB components: the wires and profiles—will be done on 

what are essentially enlarged tracks on an inner or outer 

layer. 

This arrangement provides a great deal of flexibility 

in stack-up configuration. Profiles/wires are bonded to 

a routing track, and thermal dissipation planes can be 

located on the same layer or on a facing layer co-axial 

with the MiB track. Facing layer thermal planes have been 

seen to improve thermal performance as shown in Fig. 3. 

Design guidelines include ampacity tables based on actual 

thermographic observations of different layup configurations. 

As shown in Fig. 4, the DWPCB type is suited for 

“medium” power applications with nominal currents up 

to about 140 A (40 °C temp rise). In combination with 

thermal vias or inlays this value can run to over 300 A 

depending on duty cycle. Other MiB types covered in 

BPA’s report are designed for high current applications 

(250 - 1000 A) — typical of Hybrid/Electric Vehicle and 

high power rectification. 

The medium current capability, heat-spreading characteristics, 

and design versatility of the DWPCB type make 

72.8 

102.9 

126.1 

145.6 

162.7 

178.3 

Fig. 4. Temperature increase for various profile cross-sections and current in 

Amperes (source: Häusermann GmbH). 

Cu – Profile 

12 mm 

it a cost-effective alternative to logic 

boards, busbars, and cable in an expanding 

range of applications. 

ELECTROMOBILITY POWERTRAIN 

The Lithium-ion battery pack shown in 

Fig. 5 provides a mean value of 100 A 

with peaks to 300 A for a light electric 

vehicle. Both weight and size are critical 

in this application as any increase in 

mass directly affects vehicle range. To 

save both, a battery management system 

consisting of control logic and eight 

DirectFET power devices is mounted 

right on the front of the battery pack. A 

DWPCB solution proved ideal for integrating 

the control logic with the power 

section, reducing two pcbs to one and 

eliminating the associated connectors and 

cabling. 

The design challenge involved rout- 

ing 100 / 300 A through an FR-4 circuit board to the 

DirectFETs and out to the load. Häusermann’s versatile 

DWPCB technology enabled several different options to 

be considered, ranging from single-layer power distribution 

to a multi-level design with integrated thermal vias 

and dissipation planes. 

The most cost-effective design is shown in Fig. 6. In 

this layout, the profiles have been bonded to an inner core 


97.6 

138.1 

169.1 

195.3 

218.4 

239.2 

MOSFETS 

Fig. 5. Fraunhofer IIS Battery Management System/Controller (source: 

Häusermann GmbH)

Fig. 6. MiB layout for DirectFET package using HSMtec. 

consisting of High Tg FR-4 clad with 2 oz. (70μ) copper. 

Since the profiles are all on the same reference plane, 

setup and run time for the bonding equipment is optimized 

and the subsequent mass-lamination process has a 

less complex embedding task compared to a design where 

the profiles are on several different layers. The embedded 

profiles also free up real estate on the surface of the board: 

the area needed for the high-current tracks is confined to 

the via arrays used to access the embedded profiles,while 

the wide tracking necessary to support the profiles is confined 

to the internal layer. 

MICROVIAS 

The high current profiles are accessed by arrays of laser 

microvias. This technique is growing in popularity in a 

number of MiB types, because although laser vias are 

typically 100μ in diameter or less, what counts in an MiB 

application is the length of the conductive element (in this 

case, depth of the via) and cross-sectional area of copper 

available as a thermal pathway. 

The copper area can be determined by the difference 

in surface area between the two circles formed by the via 

drilled diameter and inside plated diameter. The thermal 

resistance of the via array is therefore determined by: 

R Ѳ array = l / k × (N vias × {π × [(D 1 /2) 2 - (D 2 /2) 2 ]}) (2) 

Where: 

l = Depth of via 

k = Conductivity of copper (approx. 380 W/m-K) 

D 1 = Drilled diameter 

D 2 = Finished (plated) 

hole diameter 

N vias = Number of vias 

in the array 

And, since laser 

microvias are very small 

and are drilled at effective 

hit rates well in 

excess of 10,000 per 

minute (for CO2 “copper 

direct” process), 

a lot of them can be 

placed in a thermal pad. 

An array of 1296 microvias 

in a 100 mm2 Fig. 7. The finished board: DirectFET footprint 

(source: Häusermann GmbH). 

thermal 

pad will present a thermal resistance of about 0.01 

W/°C: about the same as the solder joint used to bond the 

device’s thermal slug to the pad! 

In addition to the microvias used to access the buried 

profiles, the design increases the dissipation area available 

to the device by running thermal vias down to bottomside 

dissipation planes. Here, the fabrication challenge entails 

drilling bulk copper, which involves chiploads less than 

one-third those of FR-4. However, thermal vias are basically 

heat pipes, so any nailheading that occurs at the layer 

3 joint in this design will not compromise electrical performance. 

The objective is to get a clean hole and homogeneous 

copper deposit for thermal transfer. Thermal via 

arrays can be very effective ways to get heat down through 

the board to a backside dissipation plane: the conductivity 

of a typical array is between 20 – 30 W/m-K which is over 

100X that of FR-4. 

Managing the power tracks inside the board leaves 

room for signal and control wiring on the outer layer, and 

the gate signals for the parallel FETs are run on the surface 

of the board. Typical dimensions for a layout of this kind 

are shown in Fig. 6 as well as the positioning of the profiles 

on the etched core carrier tracks. Gaps between the 

profiles have been expanded for clarity: the gap volume 

represents a variation of less than 0.1% of the total conductor 

volume. The result of this design is the clean footprint 

shown in Fig. 7. 

MIB COST 

Faced with a solution presenting this level of innovation 

and simplicity, the first question is almost certainly “but 

how much does it cost?” While the applications requiring 

this solution are driven primarily by reliability and 

performance (electromobility lifetimes range from 15 to 

25 years, and improvements in functional density mean 

decreases in weight and volume with corresponding 

increases in operational autonomy), the cost equation is 


THERMALmanagment 

zero sum, i.e., an alternative technology either has to 

come in at parity and be ready to follow the same cost 

reduction trajectory as the assembly it replaces, or offer an 

order of magnitude improvement: in cost, performance, 

reliability, or a combination of all three. 

The design example presented above essentially 

accomplished two objectives: 

• it enabled the use of the latest generation of isometric 

power packages, in the process reducing board real 

estate necessary for thermal and current management 

by some 30% 

• it enabled a reduction in volume and weight through a 

combination of reduced power board area, elimination 

of the controller daughterboard, replacement of the 

vertical plate heat sinks required by PIH (pin-in-hole) 

TO devices with smaller finned assemblies and chassis 

standoffs, and replacement of high current cabling with 

direct connections to the power bus. 

Considering only the reduction in parts count and 

assembly costs, the MiB solution provided a savings of 

about 13% compared to the initial design. But the potential 

savings could be greater when taking into account the 

additional benefits of weight and volume reduction. The 

cost savings obtained from reduced weight and volume is 

the subject of ongoing analysis. 

The result of this design has been a printed circuit solution 

to the challenges of heat dissipation and power distribution 

typical of medium power applications using the 

latest SMT packages. The surface mount nature of these 

devices means the board must be capable of providing low 

resistance thermal as well as electrical pathways. This 

design exemplifies how conventional printed circuit technologies, 

including laser and mechanically drilled holes , 

may be combined in innovative buildups to meet the 

challenges of Digital Power. 

*NOTE* 

Information for the Metal In The Board article was derived from a February 

2013 BPA Consulting report on “Metal In the Board-Opportunities 

for Printed Circuit Boards Providing Enhanced Thermal and Power 

Management.” In the report, BPA (www.bpaconsulting.com) take the first 

comprehensive look at what Digital Power means for printed circuits. In over 

250 pages including 360 illustrations, charts, and tables, the report presents 

the fundamentals of thermal and power management for PCBs, identifying 

and profiling a number of emerging board level solutions and applications. 

The report includes both technology and business analysis as well as international 

market forecasts by technology, application, and region. 


DESIGNfeature 

CHRISTOPHE BASSO, Product Engineering Director, ON Semiconductor, Toulouse, France 



“Designing Control Loops for 

Linear and Switching Power 

Supplies” is the latest book 

from Christophe Basso, a 

past contributer to Power 

Electronics. This book focuses 

on what engineers really 

need to know to compensate 

or stabilize a given control 

system. This article contains 

excerpts from the section of 

the book covering stability 

criteria. 

+ (s) 

Vin (s) H(s) 

V out (s) 

– 

G(s) 

Fig. 1. An oscillator is actually a control system where 

the error signal does not oppose the output signal 

variations. 

In the electronic field, an oscillator is a circuit capable of producing a selfsustained 

sinusoidal signal. In a lot of configurations, cranking up the oscillator 

involves the noise level inherent to the adopted electronic circuit. As the 

noise level grows at power-up, oscillations are started and self-sustained. This 

kind of circuit can be formed by assembling blocks such as those appearing 

in Fig. 1. As you can see, the configuration looks very similar to that of our 

control system arrangement. 

In our example, the excitation input is not the noise but a voltage level, V in , 

injected as the input variable to crank the oscillator. The direct path is made of 

the transfer function H(s), while the return path consists of the block G(s). To 

analyze the system, let us write its transfer function by expressing the output 

voltage versus the input variable: 

If we expand this formula and factor V out (s), we have 

The transfer function of such a system is therefore 

In this expression, the product G(s)H(s) is called the loop gain, also noted 

T(s).To transform our system into a self-sustained oscillator, an output signal 

must exist even if the input signal has disappeared. To satisfy such a goal, the 

following condition must be met: 

To verify this equation in which V in disappears, the quotient must go to 

infinity. The condition of the quotient to go to infinity is that its characteristic 

equation, D(s), equals zero: 

www.powerelectronics.com June 2013 | Power Electronics Technology 13 

 

1 + G(s)H(s) = 0 (5) 

To meet this condition, the term G(s)H(s) must equal -1. Otherwise 

stated, the magnitude of the loop gain must be 1 and it sign should change to 

minus. A sign change with a sinusoidal signal is simply a 180° phase reversal. 

These two conditions can be mathematically noted as follows:

POWER CONTROLloops 

–180 –30.0 

–360 

° dB 

360 

180 

0 

60.0 

30.0 

0 

–60.0 

|T (f)| 

T (f) 

100 1k 10k 100k 

Frequency (Hz) 

|G(s)H(s)| = 1 (6) 

|T (100 kHz)| = 0 dB 

T T (100 kHz) = –180° –180° 

argG(s)H(s) = –180º (7) 

1M 10M 

Fig. 2. Conditions for oscillation can be illustrated either in a Bode diagram or in a Nyquist plot. 

When these two expressions are exactly satisfied, we 

have conditions for steady-state oscillations. This is the 

so-called Barkhausen criterion, expressed in 1921 by the 

eponymous German physicist. Practically speaking, in a 

control loop system, it means that the correction signal 

no longer opposes the output but returns in phase with 

the exact same amplitude as the excitation signal. In 

dB ° 

40 

20 

0 

–20 

–40 

180 

90 

0 

–90 

–180 

|T (f)| 

T (f) 

φ m = 90° 

Crossover frequency frequency 

f c = 6.5 kHz 

10 100 

1k 10k 100k 


 

a Bode plot, (3.6) and (3.7) would imply a loop gain 

curve crossing the 0-dB axis and affected by a 180° phase 

lag right at this point. In a Nyquist analysis, where the 

imaginary and real portions of the loop gain are plotted 

versus frequency, this point corresponds to the coordinates 

-1, j0. Fig. 2 displays these two curves where conditions 

for oscillation are met. Should the system slightly 

deviate from these values (e.g., temperature drift, gain 

change), output oscillations would either exponentially 

decrease to zero or diverge in amplitude until the upper/ 

lower power supply rail is reached. In an oscillator, the 

designer strives to reduce 

as much as possible the 

gain margin so the conditions 

for oscillations are 

satisfied for a wide range 

of operating conditions. 

3.2 STABILITY CRITERIA 

You understand that our 

goal with a control system 

is not to build an 

oscillator. We want a 

control system featuring 

speed, precision, and an 

oscillation-free response. 

We must therefore keep 

away from a configuration 

where conditions for 

oscillation or divergence 

are met. One way is to 

limit the frequency range 


 

 

 

 

 

T T (s) = –180° 

1M 

ω 

–1,j0 

GM = 45 dB 

Fig. 3. In this example, the 0-dB crossover point is located at 6.5 kHz, where the total phase lag offers a phase margin of 90°. 

Tm T (ω) 

Re T (ω)

within which our system will react. By definition, the 

frequency range, or the bandwidth, corresponds to a 

frequency where the closed-loop transmission path from 

the input to the output drops by 3 dB. The bandwidth 

of a closed-loop system can be seen as a frequency range 

where the system is said to satisfactorily respond to its 

input (i.e., follows the setpoint or efficiently rejects the 

perturbations). As we will later see, during the design 

stage, We do not directly control the closed-loop bandwidth 

but the crossover frequency f c , a parameter pertinent 

to an open-loop analysis. Both variables are often 

approximated as equal, and we will see that it is true in 

one condition only. However, they are not far away from 

each other, and both terms can be interchanged in the 

discussion. 

We have seen that the open-loop gain represents 

an important parameter for our control system. When 

gain exists (i.e. |T(s)|>1), the system works in dynamic 

closed-loop conditions and can compensate for incoming 

perturbations or react to setpoint changes. However, 

there is a limit to the system reaction: the system must 

offer gain at the frequencies involved in the perturb- 

ing signal. If the perturbations of the setpoint changes 

are too fast, the frequency content of the excitation 

signal is beyond the bandwidth of the system, implying 

the absence of gain at these frequencies: the system 

becomes slow and cannot react, operating as if the 

loop were unresponsive to varying waveforms. Is an 

infinite bandwidth system desirable then? No, because 

increasing the bandwidth is like widening the diameter 

of a funnel: you are certainly going to collect more 

information and react faster to incoming perturbations, 

but the system will also accept spurious signals such as 

noise and parasitic data, self-produced by the converter 

in some cases (the output ripple in switching power 

supplies, for instance). For this reason, it is mandatory 

to limit the bandwidth to what your application really 

requires. Adopting too wide a bandwidth would be 

detrimental to the noise immunity of the system (e.g. 

its robustness to external parasitic signals). 

LIMITING BANDWIDTH 

How do we limit the bandwidth of a control system? 

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“Time ago, when amplifiers or servomechanisms were driven by 

vacuum tubes-based circuits, warm-up times during the power-on 

sequence could induce large loop gain variations.” 

point as well as the phase margin. We know by experience 

that the elements constitutive of the converter will 

exhibit variations along the product life cycle. These 

variations can be linked to natural production spreads 

(for instance, resistors or capacitors affected by lot-tolot 

tolerance). The converter environmental operating 

conditions also have an impact on components. Among 

these variables, temperature plays an important role 

and affects passive or active component parameters. It 

can be capacitors or inductors equivalent series resistors 

(ESR), the optocoupler current transfer ratio (CTR), 

or the beta of bipolar transistors for instance. These 

variations impact the loop gain by shifting it up or down 

depending on the affected parameters. 

If the gain curve undergoes a shift, the 0-dB cross- 

over frequency will transition to a new value imposing a 

different bandwidth to the converter. How can the converter 

stability be affected under these changes? Well, if 

the new crossover tales place at a point where the phase 

margin is weak, you may degrade the transient response 

so that the overshoot is no longer acceptable. It is thus 

your responsibility, as a designer, to ensure that these 

dispersions do not suddenly increase the gain at a frequency 

where you approach the -180° limit. You need 

sufficient gain margin as defined by 


POWER CONTROLloops 

dB ° 

40.0 

20.0 

0 

–20.0 

–40.0 

180 

|T (f)| 

90.0 

0 

–90.0 

–180 

T (f) 

–2 dB 

–10 dB 

φ m 45° 

φ φm 0° m 

0° 

Where corresponds to the frequency point where is 

exactly -180° or radians (1 MHz in Fig. 3). 

Fig. 4. portrays typical gain variations of ±10 dB due to 

production spreads in the selected components. It brings 

the crossover frequency from 1.5 kHz to 30 kHz. In this 

area, the phase margin changes from 70° to 45°, safe 

numbers according to theory. What is the worst case? 

It is when the new crossover frequency occurs where 

the total phase lag is 180°, matching the conditions for 

oscillations. This condition occurs at 1 MHz, implying a 

positive gain change of 35 dB. 

LARGE GAINS UNLIKELY 

Fortunately, deviations of 35 dB are unlikely to happen 

in modern electronics circuits. Time ago, when 

amplifiers or servomechanisms were driven by vacuum 

tubes-based circuits, warm-up times during the poweron 

sequence could induce large loop gain variations. 

Gain provisions were thus necessary to reject a second 

point where the stability could be in danger. This gain 

margin, identified on the loop gain curve at the frequency 

where the total phase lag reaches -180°, is noted 

GM in Fig. 3. In modern electronic circuits, gain margins 

beyond 10 dB are usually enough, unless your loop gain 

exhibits extreme sensitivity to an external parameter. 

Another example of gain shift appears in Fig. 5. It 

shows another compensated converter exhibiting a phase 

margin of 80° at 10 kHz. Based on what we discussed, 

we know that gain changes can occur, inducing ups or 

downs on the gain curve. In our example, we can identify 

an area around 2 kHz where the phase margin is as 

Crossover frequency 

f 

c = 10 6.5 kHz 

c 

 

φ = 50° m 

10 100 

1k 


10k 100k 

Fig. 6. The phase lags beyond 180° but in an area where the gain is larger than 1. This is not a problem, 

and the response is acceptable. 

small as 18°. If a gain decrease of 20-25 

dB occurs, you can end up with a control 

system showing a dangerously low 

phase margin around 2 kHz. It would 

lead to an oscillatory response, perhaps 

exceeding the overshoot specifications. 

This kind of system is told to be conditionally 

stable. Fortunately, as already 

said, a 25-dB variation of gain is unusual 

and such a system can be considered 

robust with this gain margin. However, 

I have seen design cases where the 

end user (your customer) clearly stated 

in the specifications that conditional 

designs were not acceptable, asking for 

a phase margin greater than 60° at all 

points below the crossover frequency. 

In this case, it becomes mandatory to 

compensate the converter so that no 

region of reduced phase margins below 

crossover ever exist whatever the operating 

conditions are. 

STABLE OR UNSTABLE? 

It is often believed that a system where the phase dips 

below -180° before crossover is an unstable system. Such 

a response appears in Fig. 6. The phase curve quickly 

drops after 1 kHz and passes the -180° limit at 1.5 kHz 

for a few kilohertz. 

It then goes up again to offer a phase margin of 50° 

at 10 kHz. Yes, this system is stable simply because we 

do not satisfy (3.7) at 0 dB. Remember, to cancel the 

denominator of (3.3), you must have the gain magnitude 

exactly equal to 1 and a phase lag of 180° or beyond. In 

the graph, we can see that this condition is not satisfied 

at any point in the picture. However, it is worth noting 

that the loop is highly conditional. Should the gain 

reduce by a few decibels and your phase margin becomes 

less than 45°. Another 10-dB decrease and you enter a 

dangerous area of zero phase margin where, this time, 

the oscillation criteria would be met. 

Note: This article is reprinted from the book “Designing 

Control Loops for Linear and Switching Power Supplies: A 

Tutorial Guide,” (c) 2012, with permission of the publisher, 

Artech House, Inc., Boston. The subjects in the book include: 

Basics of Loop Control, Transfer Functions, Stability Criteria 

of a Control System, Compensation, Operational 

Amplifier-Based Compensation, Operational Transconductance 

Amplifier-Based Compensation, TL-431-Based Compensation, 

Shunt Regulator-Based Compensation, Measurements 

and Design Examples. The book can be purchased at 

ArtechHouse.com, Amazon.com, or BN.com 


DESIGNfeature 

MICHAEL DE ROOIJ, Ph.D., Executive Director of Applications Engineering, Efficient Power Conversion Corporation 

JOHAN STRYDOM, Ph.D., Vice President, Applications, Efficient Power Conversion Corporation 

eGaN ® FET- Silicon Power Shoot- 

Out: Small Signal RF Performance 

In this eGaN FET-silicon power 

shoot-out series article, we 

examine RF performance 

using the 200 V EPC2012 [3] 

eGaN FET as a starting point. 

The eGaN FET is optimized 

as a power-switching device 

but also exhibits good RF 

characteristics. Future eGaN 

FET parts can be optimized 

for better RF performance at 

higher frequencies. 

1621 

Gate Circuit 

Reference Plane 

Device Outline 

914 

349 

271 

135.3 

Unlike power switching FETs, RF FETs are designed to work best 

in the linear region of operation to maximize power gain and 

minimize distortion, whereas power switching devices are optimized 

for lowest R DS(ON) and gate charge [1,2,3,17,18,19,20]. Another 

significant difference between power switching and RF FETs is the 

power dissipation capability of RF devices is significantly higher 

than that of power switching devices for equivalent terminal characteristics 

to accommodate the higher power losses in the linear region. 

We will focus on RF characterization in the frequency range from 200 MHz 

through 2.5 GHz, the results of which can be used to design a pulsed power RF 

amplifer. 

RF CHARACTERIZATION 

Prior to being able to compare various RF FETs with each other they need to be 

properly characterized, which can accomplished by measuring the S-parameters 

of the FET while regarding it as a 2-port network under controlled bias conditions. 

A test fixture was designed for the EPC2012 to connect the RF signals to the FET 

and to provide the necessary S-parameter measurement reference planes from 

which the dataset would be valid. The test fixture design used a 30 mil thick Rogers 

4350 substrate [21] , chosen for its low losses at higher frequencies. This allowed the 

design to be suitable for frequencies as high as 12 GHz. Fig. 1 shows the reference 

plane design and highlights the outline of the EPC2012 device. The transmission 

lines to the device Gate and Drain were designed as microstrip transmission lines 

with 50 Ω characteristic impedance. 

271 

914 

1621 

Drain Circuit 

Reference Plane 

Fig. 1. Reference plane design for the EPC2012 eGaN FET using a 30 mil thick Rogers 

4350 substrate. 

The test fixture was also equipped with a negative 

temperature co-efficient thermistor (NTC) placed in 

close proximity to the source pad of the eGaN FET to 

provide an indication of the temperature of the copper 

in that area without affecting the RF performance. 

The EPC9903 test fixture in Fig. 2 shows the right side 

image showing the top mounted heat-sink. The EPC 

FET has a lower thermal resistance [3] from junction to 

the top side of the device, compared to the bottom side 

(soldered), and hence mounting the heat-sink to the 

back side of the device has a high impact on the power 

dissipation capability of the device. 

Due to thermal limitations of the test fixture and the 

EPC2012 device, testing of the eGaN FET was pulsed 

with a low duty cycle with the average power dissipation 

kept below 0.7 W without the heat-sink, and 5 W 

with the heat-sink and forced air cooling. The heat-sink 

used was 15 mm x 15 mm x 14.5 mm high, supplied by 


eGaNfets 

SMA 

Connector – 

Gate 

Connection 

(AC & DC) 


Connection 



Advanced Thermal Solutions [14] with thermal interface 

material from Wakefield [7] . 

RF MEASUREMENT 

The basic setup using the EPC9903 test fixture is 

shown in Fig. 3. Both the bias and RF signal are provided 

to the board using SMA connectors. A bias Tee [4] was 

used for the separate connection of gate/ drain bias and 

the RF signal. 

Prior to using the test fixture (EPC9903) for the 

S-parameter measurement of the EPC2012 device, it was 

calibrated using the Thru-Reflect-Line (TRL) method [10] . 

The process followed is well documented and similar to 

that described in [11] . 

With the small signal S-parameter setup complete, the 

next step was to measure the EPC2012 device at various 

bias conditions to determine the highest maximum gain 

SMA 

Connector – 

Drain 

Connection 

(AC & DC) 

Heat-Sink 

Mounting Shim 

Fig. 2. Photograph of the EPC9903 small signal RF test fixture for the EPC2012 eGaN FET (with the heat sink mounted shown in right image). 

Gate 

Bias 

To VNA 

Chnl 1 

Bias 

Tee 

Gate Bias 

Voltage 

V 

O V ~ 

5 VDC 

Drain Bias 

Voltage 

10 VDC ~ 

160 VDC 

Fig. 3: Basic test fixture schematic and RF small signal test setup. 

V 

EPC9903 

Fixture 

Board 

Drain Bias 

Current 

Heat Sink 

Protective Spacer For 

Horizontal Alignment 

bias point. This will then be used to design a class A power 

amplifier to evaluate the RF power performance of the 

EPC2012. 

Initial testing to determine the useful gain frequency 

range of the device swept the frequency from 30 MHz 

through 12 GHz under continuous wave (CW) conditions 

at a low drain bias voltage of 10 V and 300 mA. 

Subsequent testing was limited to the 200 MHz through 

2.5 GHz frequency range. Initial testing also investigated 

the influence of the heat-sink on the RF performance 

of the test fixture and the device. It was found that the 

impact of the heat-sink only became detectable at frequencies 

above 2.5 GHz and caused a resonance around 

6.5 GHz which is well above the working frequency of 

the device. 

Various drain bias conditions were then applied to 

the FET terminals from 10 V through 70 V, and from 

10’s of mA though 6 A as small sig- 

nal S-Parameter measurements were 

taken. Under these conditions the bias 

power dissipation in the FET became 

significant and therefore the device 

was pulsed for short durations while 

measurements were taken. The pulse 

width was set to approximately 20 μs 

with a repetition frequency of 50 Hz. 

For this discussion, the gate-source 

circuit will be designated as port-1 

and the drain-source circuit as port-2. 

Maximum gain at 500 MHz is shown 

in the graph (Fig. 4) as a function of 

drain bias power. The graph clearly 

shows that once the drain bias power 

exceeds 20 W there is very little 


A 

Bias 

Tee 

300 mADC 

~ 3ADC 

To VNA 

Chnl 2 

Drain 

Bias

Max Gain (dB) 

20 

18 

16 

14 

12 

10 

8 

6 

4 

2 

0 

Region of Interest 

Drain 

10 V 

20 V 

30 V 

40 V 

50 V 

60 V 

65 V 

70 V 

0 25 50 75 100 125 150 175 200 

Drain Bias Power (W) 

Fig. 4. Graph of small signal Max. Gain as function of various Drain bias powers 

at 500 MHz as measured for the EPC2012 eGaN FET. 

increase in maximum gain with further increase in drain 

bias power. It also shows a nearly constant gain with drain 

voltage beyond 15 V bias. The graph shows the useful 

drain bias power range for a class A amplifier highlighted 

by the region of interest and will be the design point for 

such an amplifier. It is important to note that, for an 

amplifier design the drain bias must have sufficient voltage 

to allow the drain to swing with maximum amplitude. 

Too high of a voltage will lead to unnecessary drain bias 

power, and too low of a voltage will reduce the 1 dB compression 

point and induce clipping. 

Fig. 5 shows a graph of maximum gain as a function of 

Max. Gain (dB) 

30 

25 

20 

15 

10 

5 

0 

100 1000 

Frequency (MHz) 

Fig. 5. Maximum gain of the eGaN FET at various drain bias conditions. 

frequency for various drain bias power conditions ranging 

from 10 W through 179 W. It should be noted that low 

drain bias power reduces the gain more above 600 MHz 

and, at very high drain bias power, gain reduces below 

400 MHz. 

Three optimal drain bias points have been identified 

from the data that will be used to evaluate the performance 

of the EPC2012 device as a class A amplifier operating 

at 500 MHz. The drain voltage is around 65 V with 

drain power of 20 W, 40 W and 80 W. The 500 MHz 

point was chosen as it yielded the highest gain frequency 

product for various devices tested. The various drain bias 

power points will be used to determine the impact on the 

1 dB compression point and drain efficiency. 

RF POWER AMPLIFIER DESIGN 

Suitable drain bias points and frequency have been 

selected for the EPC2012 device based on maximum gain 

and frequency. The S-parameters at these bias points and 

frequency that can be used in the design of an RF Power 

amplifier will be analyzed next. 

Fig. 6 shows the Smith Chart plot for the gate (S11) 

and drain (S22) reflection coefficients from 200 MHz 

through 2.5 GHz with a drain bias of 64 V and 1.275 A. A 

change in drain current has negligible impact on the input 

and output impedances. However, a reduction in drain 

voltage to below 15 V will have a significant impact on 

the input and output impedance as the COSS of the eGaN 

FET increases dramatically. This will also reduce the available 

gain as more output current is shunted internally in 

the device reducing the output voltage swing. This leads 

to non-linear behavior and must be 

accounted for in an amplifier design. 

The Smith chart plot shows that the 

EPC2012 device has low impedance 

for both the gate and drain circuits in 

the frequency region from 200 MHz 

through 2.5 GHz, and of particular 

interest at 500 MHz where both are 

capacitive. 

Based on measurement data, at 

a frequency of 500 MHz the gatesource 

impedance is 5.44 – j3.69 Ω 

and the drain-source impedance is 3.13 

– j3.08 Ω and can be used to determine 

matching networks for the device. The 

impact of bias networks, and whether 

an amplifier must be unconditionally 

stable, must also be considered prior to 

matching network design. 

A stability analysis, based on the 

Rollett Stability [22] condition where 

1 [1-5] Drain 

70 V, 10 W 

64 V, 20 W 

67 V, 43 W 

64 V, 82 W 

50 V, 102 W 

55 V, 118 W 

50 V, 167 W 

40 V, 179 W 

Peak Gain 

, indicates that 


eGaNfets 

0 

0.1 

0.2 

200 MHz 

0.1 

0.2 

500 MHz 

RF Café 

2002 

0.2 

2.5 GHz 

0.3 

the EPC2012 device is close to, but not unconditionally 

stable at 500 MHz, where = 0.722 and K = 0.673. 

Fig. 7 shows the stability circle plotted at 500 MHz, 64 

V and a drain bias of 1.275 A with the unstable regions 

highlighted. A change in drain bias current, as in the case 

of input and output impedances, has a negligible impact 

on the location and size of the stability circles. The Smith 

Chart shows that the unstable regions are small. To ensure 

unconditional stability for an amplifier, a small series 

resistance in the RF gate circuit will suffice to shift the 

impedance to the right on the Smith Chart thereby ensuring 

unconditional stability. This solution is not practical for 

the output as the power may be high and the resistor will 

dissipate a large amount of RF power, thereby reducing 

the effective gain of the amplifier. A low Q factor matching 

network consisting of two L-section networks was 

chosen for the output to allow for exploration of broadband 

amplifier performance characteristics. This design 

also tends to reduce losses in the matching network since 

smaller inductors may be used with corresponding lower 

resistive loss and is particularly useful for large transformations, 

such as 2 Ω to 50 Ω. 

Since the real part of both the gate-source and drainsource 

impedances are smaller than the characteristic 

impedance of the transmission lines (50 Ω) used to connect 

the RF signal to the eGaN FET, the impedance 

matching network will take the form shown in Fig. 8. 

The basic matching network shown in Fig. 8 has the 

following solutions [1] : 

 

 

0.4 

0.5 

0.6 

S11 – Input Reflection 

S22 – Output Reflection 

0.7 

0.8 

0.9 

1.0 

Fig. 6. Smith Chart plot of Gate (S11) and Drain (S22) reflection for the EPC2012 

over the frequency range 200 MHz through 2.5 GHz. 


Source 

Load 

Unstable 

Regions 

Fig. 7. Stability Circle plot for the EPC2012 device at 500 MHz with a drain bias 

of 64 V, 1.275 A. 

(2) 

Where: 

Z o = Characteristic impedance of the transmission line 

used to connect the RF signal to the eGaN FET. 

Z L = Gate-source or drain-source impedance 

R L = Resistance part of Z L 

X L = Reactance part of Z L 

B = Matching shunt reactance 

X = Matching series reactance 

A trombone section of the input microstrip transmission 

line can be used to tune the impedance matching network 

to the device at a specific frequency whereby a shunt 

matching component may be installed anywhere along its 

length. Using the calculated value for B (Fig. 8), which in 

this case will be a capacitor, and moving it away from the 

FET on the 50 Ω transmission line, rotates the impedance 

clockwise on the Smith Chart with the result of shifting 

the impedance and altering the frequency response of the 

matching network. This is useful when designing an amplifier 

suitable for a wide operating frequency range. 

APPLICATIONS 

S-Parameter analysis of the EPC2012 eGaN FET has 

demonstrated that at up to 635 MHz, the device exhibits 

a good gain (>10 dB). This makes it useful for several 

applications, and in particular pulsed applications. These 

applications include Magnetic Resonance Imaging (MRI) 

Low power transmit systems and cyclotron drivers. 

MRI systems operate in the frequency range from

Z o 

Z in 

j-B Z L 

Fig. 8. Suitable Matching network for the EPC2012 eGaN FET. 

42 MHz (1T systems) through 300 MHz (7T systems). 

During imaging, an RF pulse is transmitted into the subject. 

The EPC2012 has several electrical characteristics 

that make it suitable for use in MRI transmit systems. 

A magnetic susceptibility test was conducted by Case 

Western University to further determine if the EPC2012 

was also suitable for use inside the MRI magnet. Any component 

inside the magnet must have an absolute volumetric 

magnetic susceptibility value of less than 24·10 -6 . Fig. 9 

shows an MRI and photograph images from the magnetic 

susceptibility test and clearly shows that the EPC device 

has no impact on the image quality and is clearly distinguishable 

in the MRI image. A device which exceeds the 

magnetic susceptibility limits will distort the image and 

will clearly show up as either a large black spot or produce 

ripples on an image similar to a stone thrown in water. 

CYCLOTRON DRIVER 

Another application that is suitable for the EPC2012 

device is a cyclotron driver. Such systems operate over 

a wide range of frequencies and are typically application 

specific. Cyclotrons are also pulsed and, due to the 

EPC2012’s small size and high voltage rating, are ideally 

suited for these types of applications [2] . Future work may 

also explore applications such as WiFi, Bluetooth, Zigbee 

and M2M power amplifiers which all operate in pulse 

mode. 

For continuous wave (CW) applications, the EPC2012 

Photograph 

Non-Magnetic 

ATC100B 

capacitor 

j-X 

EPC2012 

MRI 

Image 

EPC1013 

GRE sequence at 1.5T: TE=10 ms, TR = 100 ms 

Fig 9. Image showing the magnetic susceptibility impact of the EPC 2012 on a 

MRI image as compared to a non-magnetic ATC 100B series capacitor. 

device would need to be in an appropriate RF package that 

is capable of dissipating large amounts of heat flux. 

Another option for the efficient removal of heat from the 

RF package is eutectic die attach of the eGaN FET back 

side silicon directly to the circuit board substrate. 

ACKNOWLEDGEMENTS 

EPC hereby acknowledges the following for their support during 

this project: 

• Modelithics Inc. for their support in the design of the test fixtures 

and measuring the small signal S-Parameters. 

• Michael Twieg and Mark Griswold, Case Western Reserve 

University - Case Center for Imaging Research, for their assistance 

with the magnetic susceptibility testing of the EPC2012 

eGaN FETs 

• The blank Smith chart courtesy of www.RFcafe.com. 

• M. Meiller, Peakgainwireless, for help with s-parameter analysis 

that lead to the matching network design 

REFERENCES 

[1] D. M. Pozar, “Microwave Engineering”, Third Edition 2005, J. Wiley ISBN 

0-471-44878-8 

[2] http://en.wikipedia.org/wiki/Cyclotron 

[3] EPC2012 datasheet, www.epc-co.com 

[4] Bias Tee 8860SFM2-12, www.aeroflex.com 

[5] S. J. Orfanidis, “Electromagnetic Waves and Antennas”, Chapter 13, http:// 

www.ece.rutgers.edu/~orfanidi/ewa/ 

[6] www.rfcafe.com 

[7] Wakefield Engineering thermal interface material P/N 173-7-1212A, http:// 

www.wakefield.com 

[8] http://en.wikipedia.org/wiki/Electronic_amplifier 

[9]R. C. Hejhall, “RF Small Signal Design Using Two-Port Parameters”, 

Motorola application note AN215A, 1993. 

[10]Engen, G.F., Hoer C.A.,“Thru-Reflect-Line: An Improved Technique for 

Calibrating the Dual Six-Port Automatic Network Analyzer,” IEEE Trans. 

Microwave Theory and Techniques, December 1979. 

[11]Agilent Network Analysis Applying the 8510 TRL Calibration for Non- 

Coaxial Measurements Product Note 8510-8A 

[12] www.microwaves101.com 

[13]http://en.wikipedia.org/wiki/Magnetic_susceptibility 

[14] ATS-54150K-C2-R0 datasheet, Advanced Thermal Solutions, www.qats. 

com 

[15]Ken Payne, “Practical RF Amplifier Design Using the Available Gain 

Procedure and the Advanced Design System EM/Circuit Co-Simulation 

Capability”, Agilent Technologies White Paper, 2008, www.agilent.com 

[16]G. Gonzales, “Microwave Transistor Amplifiers”, Second Edition 1997, 

Prentice Hall ISBN 0-13-254335-4 

[17]A. Lidow, J. Strydom, M. de Rooij, Y. Ma, “GaN Transistors for Efficient 

Power Conversion”, First Edition, ISBN 978-0-615-56925-3 

[18] J. Strydom, “eGaN® FET- Silicon Power Shoot-Out Volume 8: Envelope 

Tracking”, Power Electronics Technology, May 2012, http://powerelectronics. 

com/power_semiconductors/gan_transistors/egan-fet-silicon-power-shoot-outvolume-8-0430/ 

[19]J. Strydom, “eGaN® FET- Silicon Power Shoot-Out Volume 11: 

Optimizing FET On-Resistance”, Power Electronics Technology, Oct. 2012, 

http://powerelectronics.com/discrete-semis/gan_transistors/egan-fet-siliconpower-shoot-out-volume-11-optimizing-fet-on-resistance-1001/ 

[20] M. de Rooij, J. Strydom, “eGaN® FET- Silicon Power Shoot-Out Volume 

9: Low Power Wireless Energy Converters”, Power Electronics Technology, 

June. 2012, http://powerelectronics.com/discrete-power-semis/egan-fet-siliconshoot-out-vol-9-wireless-power-converters 

[21] Rogers 4350 material specifications, www.rogerscorp.com 

[22] J. M. Rollett, “Stability and Power-Gain Invariants of Linear Twoports”, 

IRE Transactions on Circuit Theory, Vol. 9, Issue 1, March 1962, pp 29 – 32 


DESIGNfeature 

SAM DAVIS, Power Electronics 

Back to Basics: 

Voltage Regulator ICs, Part 1 

There are two basic types 

of regulator ICs, linear and 

switch-mode. The linear 

regulator operates in the 

linear region and is always 

on, whereas basic the switchmode 

type turns on and off 

and requires a rectifier to 

produce a dc output voltage. 

Among regulators, the simplest regulator circuit is for a low-dropout 

(LDO) voltage regulator whose topology is shown in Fig. 1. As a 

linear voltage regulator, its main components are a pass transistor, 

error amplifier, voltage reference, and output power MOSFET. 

One input to the error amplifier, set by resistors R1 and R2, monitors 

a percentage of the output voltage. The other input is a stable 

voltage reference (VREF). If the output voltage increases relative 

to VREF, the error amplifier changes the pass-transistor’s output to maintain a 

constant output voltage (VOUT). 

Low dropout refers to the difference between the input and output voltages 

that allows the IC to regulate the output voltage. That is, the LDO regulates the 

output voltage until its input and output approach each other at the dropout voltage. 

Ideally, the dropout voltage should be as low as possible to minimize power 

dissipation and maximize efficiency. 

The major advantage of an LDO IC is its relatively “quiet” operation because it 

does not involve switching. In contrast, a switch-mode regulator typically operates 

between 50 kHz and 1 MHz, which can produce EMI that affects analog or RF 

circuits. LDOs with an internal power MOSFET or bipolar transistor can provide 

outputs in the 50 to 500mA range. The LDO’s low dropout voltage and low quiescent 

current make it a good fit for portable and wireless applications. 

V IN 

C IN 

EN 

Pass 

Transistor 

Shutdown 

& UVLO 

+ 

Error 

Amplifier 


– 

LDO 

V REF 

Bandgap 

Voltage 

Reference 

R1 

R2 

V OUT 

C OUT 

Ground 

Bypass 

C Bypass 

Fig. 1. An LDO’s low dropout voltage and low quiescent current make it a good fit for portable and wireless applications.

An LDO regulator’s dropout voltage determines the 

lowest usable input supply voltage. That is, although specs 

may show a broad input voltage range, the input voltage 

must be greater than the dropout voltage plus the output 

voltage. For a 200mV dropout LDO, the input voltage 

must be above 3.5V to produce a 3.3V output. 

With an LDO, the difference between input voltage and 

output voltage may be small, and the output voltage must 

be tightly regulated. Plus, transient response must be fast 

enough to handle loads that can go from zero to tens of 

amperes in nanoseconds. Further, output voltage can vary 

due to changes in input voltage, output load current, and 

temperature. Primarily, these output variations are caused 

by the effects of temperature on LDO voltage reference, 

error amplifier, and its sampling resistors (R1 and R2). 

SWITCH-MODE REGULATORS 

In many applications linear supplies have been superseded 

by switch-mode supplies. Shown in Fig. 2 is a typical isolated 

switch-mode supply. One widely used approach uses 

the on and off times of pulse-width modulation (PWM) to 

control the power switch output voltage. The ratio of on 

time to the switching period time is the duty cycle. The 

higher the duty cycle, the higher the power output from 

the power MOSFET switch. A low-pass filter connected 

to the output transformer provides a voltage proportional 

to the ON and OFF times of the PWM controller. In 

operation, a fraction of the dc output voltage feeds back to 

the error amplifier, which causes the comparator to control 

the PWM ON and OFF times. If the output voltage 

changes, the feedback adjusts the duty cycle to maintain 

the output voltage at the desired level. 

To generate the PWM signal, the error amplifier 

accepts the feedback signal input and a stable voltage 

reference to produce a output related to the difference 

V CC 

PWM 

controller 

ON 

OFF 

V OUT feedback 


Switch 

Low-Pass Filter 

VOUT Fig. 2. Switch-mode power supply turns the input dc on and off, then rectifies it to obtain a dc output. 

L1 

C1 

R1 

R2 

of the two inputs. The comparator compares the error 

amplifier’s output voltage with the ramp (sawtooth) from 

the oscillator, producing a modulated pulse width. The 

comparator output is applied to the driver, whose output 

goes to the power MOSFET. 

The inductor-capacitor low pass output filter converts 

the switched voltage from the switching transformer to a 

dc voltage. The filter is not perfect so there is always some 

residual output noise called ripple. The amount of ripple 

depends on the effectiveness of the low pass filter at the 

switching frequency. Power supply switching frequencies 

can range between 100kHz to over 1MHz. Higher switching 

frequencies allow the use of lower value inductors and 

capacitors in the output low pass filter. However, higher 

frequencies can also increase power semiconductor losses, 

which reduce power supply efficiency. 

In terms of power dissipation, the power switch is key 

component in the switch-mode power supply. The switch 

is usually a power MOSFET that operates in only two 

states - on and off. In the off state the power switch draws 

very little current and dissipates very little power. In the 

on state the power switch draws the maximum amount 

of current, but its on-resistance is low, so in most cases its 

power dissipation is minimal. In the transition from the on 

state to the off state and off to on the power switch goes 

through its linear region where it consumes some power. 

The total losses for the power switch is therefore the sum 

of the on and off state losses plus the losses in the transition 

through its linear region. 

CONVERTER ICS 

ICs for switch-mode power supplies are found either of 

two basic configurations: converter and controller ICs. 

Converter ICs provide a complete dc-dc converter in a 

single package. The only required external components are 

usually passive devices. Power switches 

may be either a bipolar or MOSFET 

device capable of handling the required 

current and power. Typically, the power 

semiconductor switch turns on and off 

at a frequency that may range from 100 

kHz to 1 MHz, depending on the IC 

type. Most power switches employ pulse 

Load width modulation to control the output 

voltage, so the duty cycle varies according 

to the desired output voltage. 

A controller IC requires an external 

power switch, either a bipolar transistor 

or power MOSFET. The controller 

circuit that employs an external power 

switch usually has higher efficiency than 

the converter with an integrated power 


VOLTAGEregulator ICs 

MOSFET because integrated 

MOSFETs have a higher onresistance 

(higher losses). 

On-resistance of an external 

power MOSFET is lower 

and the MOST usually has 

higher power output capa- V 

IN 

bility than an IC with an 

integrated MOSFET. 

For both converter and 

controller ICs, the switching 

frequency determines the 

physical size and value of 

filter inductors, capacitors, 

and transformers. The higher 

the switching frequency, 

the smaller the physical size 

and component values. To 

optimize efficiency, magnetic 

core material for the inductor and transformer should 

be consistent with the switching frequency. That is, the 

transformer/inductor core material should be chosen to 

operate efficiently at the switching frequency. 

 

FREE SAMPLES 

C1 

Switch Network 

Oscillator & Logic 

Fig. 3. An advantage of the charge pump is elimination of the magnetic fields 

and EMI that comes with an inductor or transformer. 

DC-DC converters 

accept a dc input and produce 

a dc output. They can 

be isolated or non-isolated, 

which depends on whether 

there is a direct dc path 

V from the input to the out- 

OUT 

put. An isolated converter 

(Fig. 2) employs a trans- 

C2 former to provide isolation 

between the input and 

output voltage. The nonisolated 

converter employs 

an inductor-capacitor filter 

and an optocoupler 

usually provides isolation 

between the output feedback 

and the input. For 

many applications, nonisolated 

converters are appropriate. An advantage of the 

transformer-based converter is that it has the ability to 

easily produce multiple output voltages using multiple 

secondary windings. 

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Initially, integrated power switch 

converters used bipolar power switches, 

but virtually all newer devices 

employ MOSFET power switches that 

improve efficiency. Another efficiency 

improvement is the use of integrated 

synchronous rectifiers consisting of 

power MOSFETs switches that rectify 

the power supply output and provide 

a dc output. 

Among the features found in converter 

and controller ICs: 

• Fixed or an adjustable output voltage 

• Single-ended or synchronous outputs 

• Soft-start that causes the output to come up gradually 

• Undervoltage lockout 

• Thermal shutdown 

• Overcurrent protection 

• Overvoltage protection 

CHARGE PUMP ICS 

Charge pumps are actually a different form of switching 

supply. They switch capacitors to provide dc-dc voltage 

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A major advantage 

of the charge pump 

is elimination of the 

magnetic fields and 

EMI that comes 

with an inductor or 

transformer. 

conversion using a switch network 

to charge and discharge one or more 

capacitors. The switch network toggles 

between charge and discharge states of 

the capacitors. As shown in Fig. 3, 

the “flying capacitor “ (C1) shuttles 

charge, and the “reservoir capacitor “ 

(C2) holds charge and filters the output 

voltage. The basic charge pump 

lacks regulation, which is generally 

added using either linear regulation 

or charge-pump modulation. Linear 

regulation offers the lowest output 

noise, and therefore provides better 

performance. Charge-pump modulation offers more output 

current for a given die size (or cost), because the 

regulator IC need not include a series pass transistor. 

A major advantage of the charge pump is elimination 

of the magnetic fields and EMI that comes with an inductor 

or transformer. There is one possible EMI source - the 

high charging current that flows to a “flying capacitor” 

when it connects to an input source or another capacitor 

with a different voltage. 

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VOLTAGEregulator ICs 

Gate 

Drive 

MOSFET 2 

MOSFET 1 

V CC 

MULTIPLE OUTPUT CONVERTER/REGULATOR ICS 

Multiple output controller ICs consist of two or more 

regulators in a single package. They could be two switchmode 

converters or two LDO regulators. 

An example of a dual switch-mode regulator is a dual 

current mode PWM step-down dc-dc converter with 

internal 2 A power switches, this IC operates from a 3.6 

V to 25 V input, enabling it to regulate a wide variety 

of power sources such as four-cell batteries, 5 V logic 

rails, unregulated wall transformers, lead acid batteries 

and distributed-power supplies. The two regulators share 

common circuitry including input source, voltage reference 

and oscillator, but are otherwise independent. Their 

feedback loop controls the peak current in the switch during 

each cycle. This current mode control improves loop 

dynamics and provides cycle-by-cycle current limit. 

An example of a dual-output, low-dropout voltage 

regulator IC has integrated reset, power on reset (POR) 

and power good (PG) functions. Differentiated features, 

such as accuracy, fast transient response, supervisory 

circuit (power on reset), manual reset input, and independent 

enable functions provide a complete system solution. 

These voltage regulators have extremely low noise output 

performance without using any added filter bypass capacitors 

and are designed to have a fast transient response and 

usually stable with low ESR capacitors. 

This LDO family also can feature a sleep mode; 

applying a high signal to either enable input shuts down 

Regulator 1 or Regulator 2, respectively. Putting the regulators 

in the sleep mode reduces the input current at TJ = 

25°C. Each regulator, has an internal discharge transistor 

to discharge the output capacitor when the regulator is 

turned off (disabled). 

L 

C OUT 

V OUT 

Fig. 4. The synchronous rectifier is more efficient than a Schottky diode rectifier. 

Multiple output controller ICs can also consist of two 

or more charge pump converters in a single package. They 

can be controllers that employ external power switches or 

regulators with an internal power switch. One possibility 

is a 5 V output and a 3.3 V output for processor and logic 

applications. 

For example, a typical multiple output charge pump 

controller ICs can step-down dc-dc converters that produce 

two adjustable regulated outputs from a single 2.7 V 

to 5.5 V input. The IC uses switched capacitor fractional 

conversion to achieve a typical efficiency increase of 50% 

over that of a linear regulator. No inductors are required. 

The IC has two switched capacitor charge pumps to 

step down VIN to two regulated output voltages. The two 

charge pumps operate 180° out of phase to reduce input 

ripple. Regulation is achieved by sensing each output voltage 

through an external resistor divider and modulating 

the charge pump output current based on the error signal. 

A two-phase, non-overlapping clock activates the two 

charge pumps running them out of phase from each other. 

SYNCHRONOUS RECTIFICATION 

Efficiency is an important criterion in designing dc-dc converters, 

which requires low power. These losses are caused 

by the power switch, magnetic elements, and the output 

rectifier. Reduction in power switch and magnetics losses 

require components that can operate efficiently at high 

switching frequencies. Output rectifiers can be Schottky 

diodes, but synchronous rectification (Fig. 4) consisting of 

power MOSFETS provide higher efficiency. 

MOSFETs exhibit lower forward conduction losses 

than Schottky diodes. Unlike conventional diodes that 

are self-commutating, the MOSFETs turn on and off by 

means of a gate control signal synchronized with converter 

operation. The major disadvantage of synchronous 

rectification is the additional complexity and cost associated 

with the MOSFET devices and associated control 

electronics. At low output voltages, however, the resulting 

increase in efficiency more than offsets the cost disadvantage 

in most applications. 

UPCOMING TOPICS 

There are other key regulator topologies. Next month, 

we will discuss the two basic IC topologies employed in dc 

power sources: the step-down, or buck converter and the 

step-up, or boost converter. Buck topology is a non-isolated 

power management configuration whose advantages 

are simplicity and low cost. The boost converter employs 

a switching technique that causes current to build up in an 

inductor and then stores the resulting voltage in an output 

capacitor. Multiple switching cycles build the output 

capacitor voltage so that the output voltage is higher than 

the input. 


DESIGNfeature 

ROGER ALLAN, Contributing Editor 

Intelligent Energy Optimization 

System Slashes Electric Bills 

The SP1000 series of hybrid 

energy management and 

correction systems from 

PMI guarantees and quantifies 

kilowatt-hour savings 

by reducing losses in power 

delivery, identifying and 

reducing harmonics, correcting 

for system imbalance, 

increasing equipment 

efficiency, and optimizing 

power utilization. 

Engineers at Power Metrics International (PMI) looked at age-old 

problems with a new perspective when they devised a patentpending 

optimization technique for electric power management. The 

result was PMI’s SP1000A series of hybrid energy management and 

correction system that addresses inefficiencies in power delivery and 

usage. It guarantees and quantifies kilowatt-hour savings by reducing 

losses in power delivery, identifying and reducing harmonics, correcting 

for system imbalance, increasing equipment efficiency, and optimizing power 

utilization. Reductions in kilowatt-hours of 3% to 7% have been regularly observed. 

Total electric bill reductions of 8% to 15% have been achieved. User return on 

investment (ROI) typically ranges from 15 months to two years. 

The SP1000A continuously measures current, harmonics, load-source imbalances, 

and power factor at a rate of 128 samples per cycle, then corrects for these 

parameters “on the fly.” It also stores data in five-minute increments for more than 

15 months. It was designed and tested by PMI and manufactured by MHT, which 

produces high-efficiency lighting products. Table 1 lists the SP1000A data sheet 

The circuit board of the SP1000A uses a conventional off-the-shelf triac and 

state-of-the-art, proprietary, long-life switches (rated for on and off switching times 

of millions before burnout like other types do), as well as small capacitors (Fig. 1). 

It dissipates one-tenth the power of similar products during switching. The hybrid 

system is available in three versions: 

208-V SP1000A (10 kVAR) 

208-V SP1000B (17 kVAR) 

480-V SP1000E (30 kVAR) 

The system’s SPIDER (Sensor Perfect 

Interface Data Exchange Routines) 

software supports constant power-consumption 

monitoring and management. 

It gives users a “dashboard” view of readings 

in real time in charts. Also, it highlights 

issues like motor deterioration, 

allowing users to be proactive instead 

of reactive in their maintenance (Fig. 2). 

The system makes thousands of decisions 

per minute, based on what it is 

reading through external and internal 

current transmitters. It addresses harmonics, 

voltage imbalance, temperature 

rise and fall, loss of current, and other 

facility issues while increasing motor 

efficiency and reducing the operating 

temperature. 

Fig. 1. The SP1000 hybrid energy management and correction system board (top) and its capacitors (bottom) 

fit in an electrical subpanel. The system takes up little space yet saves a lot on electric bills. 


ENERGYmanagement 

Underwriters Laboratories (UL) has 

approved the SP1000A system under Section 

508. According to PMI executive vice president 

Robert Turner, it is the only system on 

the market that can monitor, record, and 

quantify true kilowatt/kilowatt-hour savings 

at this price point. For more information 

on UL requirements, see the box “UL 508- 

Industrial Control Equipment”. 

The SP1000A also monitors each leg of 

a three-phase panel independently from the 

other two legs and then applies capacitance 

to each leg based on the load characteristics 

and arbitration between all governing 

parameters, according to the system’s inventor 

Hamid Pishdadian. 

Improved PFC also increases internal electrical system capacity, since unconnected 

power increases losses in the electrical distribution system and limits 

capacity for expansion. The voltage drop at the point of use also is improved. 

Voltages below equipment rating reduce efficiency, increase current, and reduce 

the motor’s starting torque. 

TABLE 1. SP1000A DATA SHEET 

Model SP1000 Family – Series – 1000 – SP1000A 

Phase Configuration 

(120/208 Wye-3Ø) L1 (Blue) L3(Black) Neutral (White) Ground 

(Green) 

Maximum Line 120/208 Volts 

Total Capacitance rated up to 180 μF per phase 

Monitoring Capabilities Metering for monitoring PF, I, V, KVAR, and THD 

KVAR Performance Range 0.05 to 3.0 per phase 

Frequency 50 – 60 Hertz 

Typical Protection 

(Not evaluated by UL LLC) 

Electrical Storms, Lightning Activity and Power Utility Spikes 

(Units are not intended to take the place of designed Power Surge 

Equipment) 

Protected 10,000 AFC per UL-810 

Lightning Stroke Protection 

(Not evaluated by UL LLC) 

Fig. 2. An hourly view of the SP1000 hybrid energy management and correction system provides a 

view of how a facility’s current levels (red) correspond with the power consumed in kilowatts (blue). 

Metal Oxide Varistors – with Transient voltage Suppressors on separate 

MOV board for internal use only. 

Circuit Breaker Required 

30 A - 3Ø Breaker or NEC approved isolation device (External to 

unit) 

Low Losses 1.2 watt per KVAR 

Dissipation Factor 0.1% at 60 Hz and 25 °C, 1% at1,000 Hz and 25 °C 

Insulation Resistance 500 MΩ per μF 

Human Protection 

Capacitor terminals isolated from human contact. All high voltage 

shielded from contact. 

Operating Temperature Range 

Capactiors: -16 °C to +70 °C (Unit rating ambient temperature at 

50 °C) 

Dimensions (LxHxD) 16” x 12” x 16” Metal Enclosure UL Listed 

Operating Life 

60,000 hours with >94% survival (Capacitors switched by electronics 

in separate bank configuration) 

General Enclosure UL Listed NEMA-1 Enclosure 16 Ga. H.R.P. & O. (1.4 mm) 

Wire Rating: 600 Volts THHN – Gasoline & Oil Resistant 11 

Wire Gauge THHN – 10 Gauge wire 

Unit Weight 30 lbs. 

UL 508 - INDUSTRIAL 

CONTROL EQUIPMENT 

REQUIREMENTS cover industrial control 

devices, and devices accessory thereto, 

for starting, stopping, regulating, controlling, 

or protecting electric motors. These 

requirements also cover industrial control 

devices or systems that store or process 

information and are provided with an output 

motor control function(s). This equipment 

is for use in ordinary locations in 

accordance with the National Electrical 

Code, NFPA 70. 

These requirements cover devices 

rated 1500 volts or less. Industrial control 

equipment covered by these requirements 

is intended for use in an ambient temperature 

of 0 - 40°C (32 - 104°F), unless 

specifically indicated for use in other 

conditions. 

A product that contains features, characteristics, 

components, materials, or systems 

new or different from those covered 

by the requirements in this standard, and 

that involves a risk of fire or of electric 

shock or injury to persons shall be evaluated 

using appropriate additional component 

and end-product requirements to maintain 

the safety level as originally anticipated by 

the standard. A product whose features, 

characteristics, components, materials, or 

systems conflict with specific provisions 

of this standard does not comply with this 

standard. Revised shall be proposed and 

adopted in conformance with the methods 

employed for development, revision, and 

implementation of this standard. 


PETinnovations 

RAJAKRISHNAN RADJASSAMY, Texas Instruments 

Gas Gauge IC Monitors Lead-Acid Battery 

State-Of-Health, State-Of-Charge 

THE bq34z110 gas gauge IC from Texas 

Instruments provides accurate operating data for 

lead- acid batteries, a capability not available from 

other sources. This 14-pin IC (Fig. 1) is the industry’s 

only scalable gas gauge device that supports multi-cell 

lead-acid battery packs with battery voltages of 4 V, 12 V, 

24 V, 48 V and higher. 

Until recently, it has been difficult to 

accurately measure and report a lead-acid 

battery’s current capacity. This could 

frustrate the end user, and require 

more batteries to ensure an adequate 

output load. The bq34z110 

solves this problem by employing 

T.I.’s proprietary Impedance Track 

algorithm that uses voltage, current, 

temperature measurements, and battery characteristics, 

to assess battery state-of-charge within 1% error 

over normal operating conditions. 

Fig. 2 shows a typical bq34z110 circuit implementation. 

The bq34z110 gauge continuously informs the user about 

a lead-acid battery’s state-of-health and state-of-charge and 

maintains up to a 95-percent accurate capacity measurement 

for the battery’s entire life. This information prevenqts 

premature shutdown and increases longevity of the battery 

and end equipment. An associated host processor can 

interrogate the gas gauge to obtain cell information, such as 

remaining capacity, full charge capacity, and average current. 

Table 1 lists the recommended operating conditions. 

Standard Commands in the bq34z110 access battery 

information, with additional capabilities provided by an 

Extended Commands set. Both sets of commands are used 

to read and write information contained within the IC’s 

control and status registers, as well as its data flash locations. 

Commands are sent from the host to the fuel gauge 

IC using HDQ or I2C serial communications engines that 

can execute during application development, pack manufacture, 

or end-equipment operation. 

The 2-byte Standard Commands consist of two data 

bytes. Two consecutive HDQ or I2C transmissions must 

be executed to initiate the command function and to read 

or write the corresponding two bytes of data. Also, two 

block commands are available to read the manufacturer 

name and device chemistry. 

Also included in the bq34z110 are 32 bytes of userprogrammable, 

non-volatile data flash memory, accessible 

via its data flash interface. The data flash memory contains 

initialization, default, cell status, calibration, configuration, 

and user information. Cell information is stored in 

this non-volatile flash memory. Many of these data 

flash locations are accessible during application development 

and pack manufacture. They cannot, generally, be 

accessed directly during end-equipment operation. 

Access to these locations is achieved 

by either use of companion evaluation 

software, through individual commands, 

or through a sequence of data-flash-access 

commands. 

Most data flash locations, however, can only 

accessible in UNSEALED mode by 

Fig 1. bq34z110 is a use of the evaluation software or 

scalable power management by data flash block transfers. These 

device that supports multi- locations should be optimized and/ 

cell lead-acid battery packs. 

or fixed during the development 

and manufacture processes. They 

become part of a Golden Image File and can then be written 

to multiple battery packs. Once established, the values 

generally remain unchanged or get updated by the gauge 

algorithm during end equipment operation. 

Two security modes control data flash access permissions: 

• Public Access refers to those data flash locations that are 

accessible to the user. 

• Private Access refers to reserved data flash locations used 

by the bq34z110. 

FUEL GAUGING 

The bq34z110 measures cell voltage, temperature, and 

current to determine the battery state of charge (SOC) 

based on the Impedance Track algorithm. It monitors 

charge and discharge activity by sensing the voltage across 



a resistor (5 mΩ to 20 mΩ , typical) 

between the SRP and SRN pins 

and in-series with the cell. By integrating 

charge passing through the 

battery, the cell’s SOC is adjusted 

during battery charge or discharge. 

When an application’s load is 

applied, cell impedance is measured 

by comparing its Open 

Circuit Voltage (OCV) with its 

measured voltage under loading 

conditions. The total battery capacity 

is found by comparing states of 

charge before and after applying 

the load with the amount of charge 

passed. When an application load 

is applied, the impedance of the 

cell is measured by comparing the 

OCV obtained from a predefined 

function for present SOC with the 

measured voltage under load. Measurements of OCV and 

charge integration determine chemical state of charge and 

Chemical Capacity (Qmax). 

The bq34z110 can use an NTC thermistor (default is 

Semitec 103AT or Mitsubishi BN35-3H103FB-50) for 

temperature measurement, or can also be configured to 

use its internal temperature sensor. It uses temperature to 

monitor the battery-pack environment, which is used for 

fuel gauging and cell protection functionality. 

To minimize power consumption, the bq34z110 has 

three power modes: NORMAL, SLEEP, and FULL 

SLEEP. The bq34z110 passes automatically between 

these modes, depending upon the occurrence of specific 

events. Multiple modes are available for configuring from 

one to 16 LEDs as an indicator of remaining state of 

charge. More than four LEDs require the use of one or 

two inexpensive SN74HC164 shift register expanders. A 

SHA-1 or HMAC-based battery pack authentication feature 

is also implemented on the bq34z110. When the IC 

is in UNSEALED mode, authentication keys can be (re) 

assigned. Alternatively, keys can also be programmed permanently 

in secure memory. A scratch pad area receives 

challenge information from a host and also to export 

SHA-1/HMAC encrypted responses. 

The bq34z110 has two flags accessed by the Flags() 

function that warns when the battery’s SOC has fallen to 

critical levels. When RemainingCapacity() falls below the 

first capacity threshold, specified in SOC1 Set Threshold, 

the [SOC1] (State of Charge Initial) flag is set. The 

flag is cleared once RemainingCapacity() rises above 

SOC1 Clear Threshold. All units are in mAh. When 

RemainingCapacity() falls below the second capacity 

TABLE 1. RECOMMENDED OPERATING CONDITIONS 

FUNCTION PARAMETER CONDITIONS MIN TYP MAX UNIT 

V REGIN 

C REGIN 

C LDO25 

I CC 

I SLP 

I SLP+ 

Supply Voltage 

External input capacitor 

for internal LDO between 

REGIN and VSS 

External output capacitor 

for internal LDO between 

VCC and VSS 

NORMAL operating mode 

current 

SLEEP operating mode 

current 

FULL SLEEP operating 

mode current 

No operating restrictions 2.7 4.5 V 

No Flash writes 2.45 2.7 V 

Normal capacitor values 

specified. Recommend a 

10% ceramic X5R type 

capacitor located close to 

the device. 

Gas gauge in 

NORMAL mode 

ILOAD = Sleep current 

Gas gauge in SLEEP mode 


Gas gauge in 

FULL SLEEP mode 


TA = 25°C, C LDO25 = 1μF, V REGIN = 3.6 V (unless otherwise noted) 

0.1 μF 

0.47 1 μF 

140 μA 

64 μA 

19 μA 

threshold, SOCF Set Threshold, the [SOCF] (State of 

Charge Final) flag is set, serving as a final discharge warning. 

If SOCF Set Threshold = –1, the flag is inoperative 

during discharge. Similarly, when RemainingCapacity() 

rises above SOCF Clear Threshold and the [SOCF] flag 

has already been set, the [SOCF] flag is cleared. All units 

are in mAh. 

The bq34z110 has two additional flags accessed by 

the Flags() function that warn of internal battery conditions. 

The fuel gauge monitors the cell voltage during 

relaxed conditions to determine if an internal short has 

been detected. When this condition occurs, [ISD] will 

be set. The bq34z110 also has the capability of detecting 

when a tab has been disconnected in a 2-cell parallel 

system by actively monitoring the state of health. When 

this condition occurs, [TDD] is set. Lead-Acid gauging in 

the charge direction makes use of four Charge Efficiency 

factors to correct for energy lost due to heat. Lead-Acid 

charge efficiency is not linear throughout the charging 

process, as it drops with increasing state of charge. 

X10 MODE 

The bq34z110 supports high current and high capacity 

batteries above 32.76 Amperes and 32.76 Ampere-Hours 

by switching to a times-ten mode where currents and 

capacities are internally handled correctly, but various 

reported units and configuration quantities are rescaled to 

tens of milliamps and tens of milliamp-hours. The need 

for this is due to the standardization of a two byte data 

command having a maximum representation of ±32767. 

When the X10 bit (Bit 7) is set in the Pack Configuration 

register, all of the mAh, cWh, and mWh settings will take 


on a value of ten times normal. When this bit is set, the 

actual units for all capacity and energy parameters will 

be 10 mAh or Wh. This includes reporting of Remaining 

Capacity. This bit will also be used to rescale the current 

reporting to 10 times normal, up to ±327 A. The actual 

resolution then becomes 10 mA. 

It is important to know that setting the X10 flag does 

not actually change anything in the operation of the 

gauge. It serves as a notice to the host that the various 

reported values should be reinterpreted ten times higher. 

X10 Current measurement is achieved by calibrating the 

current gain to a value X10 lower than actually applied. 

Because the flag has no actual effect, it can be used to 

represent other scaling values. 

VOLTAGE DIVISION AND CALIBRATION 

The bq34z110 is shipped with factory configuration for 

the default case of 2-series lead-acid cell. This can be 

changed by setting the number of series cells in the data 

flash configuration section. 

Multi-cell applications, with voltages up to 65535 

mV may be gauged by using the appropriate input 

scaling resistors such that the maximum battery voltage, 

under all conditions, appears at the BAT input as 

approximately 900 mV. The actual gain function is 

determined by a calibration process and the resulting 

PACK+ 

I 

PROG 

2 C 

HDQ COMM 

ALERT 

PACK– 

CE 

REGIN 

P1 

P2 

P3/DAT 

P4/CLK 

P5/HDQ 

optional to reduce divider power consumption 

TYPICAL IMPLEMENTATION 

BAT 

VEN 

REG 25 

P6/TS 

SRP 

SRN 

VSS 

Fig 2. Typical implementation of bq34z110 for lead-acid battery. 

voltage calibration factor is stored in the data flash location 

Voltage Divider. 

For two-cell applications, an external divider network 

is not required. Inside the IC, behind the BAT pin is a 

nominal 5:1 voltage divider with 88 KΩ in the top leg 

and 22 KΩ in the bottom leg. This internal divider network 

is enabled by clearing the VOLTSEL bit in the Pack 

Configuration register. This ratio is optimum for directly 

measuring a dual cell lead-acid cell where charge voltage 

is limited to 5 V max. 

For higher voltage applications, an external resistor 

divider network should be implemented as per the 

reference designs in this document. The quality of the 

divider resistors is very important to avoid gauging errors 

over time and temperature. It is recommended to use 

0.1% resistors with 25 ppm temperature coefficient. 

Alternately, a matched network could be used that tracks 

its dividing ratio with temperature and age due to the 

similar geometry of each element. 

AUTO CALIBRATION 

The bq34z110 provides an auto calibration feature that 

will measure the voltage offset error across SRP and SRN 

from time-to-time as operating conditions change. It subtracts 

the resulting offset error from normal sense resistor 

voltage, VSR, for maximum measurement accuracy. 

The gas gauge performs 

a single offset 

calibration when (1) the 

interface lines stay low 

for a short time. The 

bq34z110 can act as a 

SHA-1 and HMAC 

authentication slave by 

using its internal engine. 

Sending a 160-bit SHA-1 

challenge message to the 

bq34z110 causes the IC 

to return a 160-bit digest, 

based upon the challenge 

message and hidden 

plain-text authentication 

keys. When this digest 

matches an identical one, 

generated by a host or 

dedicated authentication 

master (operating on the 

same challenge message 

and using the same plain 

text keys), the authentication 

process is successful. 

www.powerelectronics.com June 2013 | Power Electronics Technology 33 

** 

** 

n Series Cells 

Sense 

Resistor


EVALUATION MODULE 

The bq34z110EVM evaluation module (Fig. 

3) is a complete evaluation system for the 

bq34z110 wide-range fuel gauge for lead-acid 

chemistries when combined with an EV2300 

USB adapter and Windows®-based PC software, 

downloadable from the TI.com website. 

The circuit module includes one bq34z110 

integrated circuit (IC) and all other components 

necessary to monitor and predict capacity 

in 2 or more series cell Lead Acid battery 

packs. The circuit module connects directly 

across the battery. 

With the EV2300 interface adapter and 

software, it is possible to read the bq34z110 

data registers, program the chip for different 

pack configurations, log cycling data for further 

evaluation, and evaluate the overall functionality 

of the bq34z110 solution under different 

charge and discharge conditions. 

600V XPT IGBTs 

Short Circuit Capability 

Rugged.Effecient.Reliable 

Part 

Number 

IXXH50N60C3D1 

IXXA50N60B3 

IXXR100N60B3H1 

IXXH30N65B4 

IXXK160N65C4 

IXXX160N65B4 

IXYN100N120C3H1 

IXYH82N120C3 

IXYK100N120C3 

V ces 

(V) 

600 

600 

600 

650 

650 

650 

1200 

1200 

1200 

I C25 

T C =25 O C 

(A) 

100 

120 

145 

65 

290 

310 

134 

160 

188 

For more parts, visit www.ixys.com 

V ce(sat) 

max 

T J =25 O C 

(V) 

2.3 

1.8 

1.8 

0.2 

2.1 

1.8 

3.5 

3.2 

3.5 

t fi 

typ 

(ns) 

42 

135 

150 

57 

30 

90 

110 

93 

110 

650V XPT Trench IGBTs 

Short Circuit Capability 

Low-on-State Voltage 

E off typ 

TJ=125 O C 

*TJ=150 O C 

(mJ) 

*0.48 

*1.2 

*2.8 

0.6 

1.3 

2.36 

3.55 

3.7 

3.55 

R thjc 

max 

( O C/W) 

0.25 

0.25 

0.31 

0.65 

0.16 

0.16 

0.18 

0.12 

0.13 

Package 

Style 

TO-247 

TO-263 

ISOPLUS247 

TO-247 

TO-264 

PLUS247 

SOT-227 

TO-247 

TO-264 

Power Inverter Switch Mode Power Supply 

Battery Charger Welding Machine 

EUROPE 

IXYS GmbH 

marcom@ixys.de 

+46 (0) 6206-503-249 

www.ixys.com 

USA 

IXYS Power 

sales@ixys.com 

+1 408-457-9042 

1200V XPT IGBTs 

High-Speed Hard-Switching 

Low Gate Drive Requirement 

ASIA 

IXYS Taiwan/IXYS Korea 

sales@ixys.com.tw 

sales@ixyskorea.com 


TO-264 

TO-220 

Fig 3. Evaluation module fuel 

gauge for lead-acid chemistries 

employs an EV2300 

USB adapter and Windows®based 

PC software. 

PLUS247 

SOT-227 

TO-247 

Uninterruptible Power Supply

NEWproducts 

■14-Bit, 4.5Msps SAR Analog-to-Digital Converter 

THE LTC2314-14 from Linear Technology is a 14-bit 4.5Msps 

successive approximation register (SAR) ADC in a tiny 8-lead 

TSOT-23 package, offering up to 90% reduction in size over 

competing solutions for space-conscious applications. This small 

SAR ADC integrates a precision 2.048 V/4.096 V reference with 

7 ppm/°C typical and guaranteed 

20 ppm/°C maximum temperature 

coefficient into an 8mm² footprint. 

The LTC2314-14 operates from a 

3 V or 5 V supply and features the 

lowest power dissipation (18 mW 

at 3 V or 31 mW at 5 V) compared 

to other competitive parts on the 

market. With its serial SPI interface, 

tiny footprint and very low power 

dissipation, the LTC2314-14 is ideal for a wide range of portable 

and space-constrained applications, including medical devices, 

communications systems and battery-operated systems. 

Also available is the LTC2315-12, a 12-bit pin- and softwarecompatible 

version of the LTC2314-14, featuring a fast 5Msps 

sample rate. The LTC2314-14 and LTC2315-12 lead a family of 

14- and 12-bit SAR ADCs featuring sample rates from 500 ksps 

■Online IGBT Product Selector and Performance Evaluator 

Tool Simplifies Device Selection and Optimization 

INTERNATIONAL RECTIFIER, IR® has enhanced its Insulated Gate Bipolar Transistor 

(IGBT) selection tool that enables design optimization in a wide range of applications 

including motor drives, uninterruptable power supplies (UPS), solar inverters, and welding. 

IR’s enhanced IGBT selection tool evaluates application conditions including bus 

voltage, load current, switching frequency, short-circuit requirements, package and 

thermal system. New features include 

customizable thermal constraint setting 

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that conveniently compare devices over 

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Located at http://mypower.irf.com/ 

IGBT, the IGBT selection tool returns 

a shortlist of products that meet or 

exceed the application parameters entered by the user. The products are ranked by 

performance with losses and junction temperature for the specified operating condition 

to help facilitate the selection process. 

IGBT selection requires evaluation of many parameters that cannot be simplified into 

a single metric, as switching losses can be traded for conduction losses. To address 

this problem, IR’s product selection tool generates a Current vs. Frequency graph that 

provides a powerful indication of the relative performance of different IGBTs. With this 

information the designer can select the most cost-effective IGBT for the application. 

International Rectifier 

El Segundo, CA 

http://www.irf.com 

to 5 Msps with 77.5 dB SNR at 14 bits and 73 dB SNR at 12 

bits resolution. Accurate DC specifications include a maximum 

INL and DNL of ±3.75 LSB and ±0.99 LSB at 14-bits, and 

±1.25 LSB and ±0.99 LSB at 12-bits, respectively. The complete 

LTC2314 SAR ADC family offers single-ended unipolar 

inputs, small 8-lead TSOT-23 

packages, precision integrated 

reference and low power dissipation 

with flexible 3V or 5V supply 

operation, along with nap and 

sleep modes for optimizing the 

power within a system. 

The LTC2314-14 and 

LTC2315-12 are available today 

in commercial, industrial and 

automotive (-40 °C to 125 °C) temperature grades. Pricing 

begins at $9.52 each for the LTC2314-14 and $5.17 each 

for the LTC2315-12 in 1,000-piece quantities. The DC1563A 

evaluation board for the LTC2314 SAR ADC family is available. 

Linear Technology 

Milpitas, CA 

http://www.linear.com 

■Miniature Through-Hole 

Power MOSFETs 

ADVANCED POWER Electronics Corp. 

(USA), launched a new series of power 

MOSFETs in the popular TO-92 throughhole 

package which is widely used in commercial 

and industrial applications where 

PCB space is at a premium. 

Ideally suited for low current applications 

such as small switch power supplies and 

load switches, the AP4002T N-channel 

enhancement-mode power MOSFETs feature 

fast switching characteristics and a low gate 

charge. BVDSS is 600 V, RDS(on) is 5 ohm, 

and ID is 400 mA. Devices also feature 

ROHS-compliant, halogen-free packaging. 

Advanced Power Electronics Corp. 

San Jose, CA 

http://www.a-powerusa.com 


NEWproducts 

■UL913 Certified Fuse For Intrinsically Safe Apparatus 

LITTELFUSE, INC. has introduced the PICO® 259-UL913 Series 

Intrinsically Safe Fuse, reportedly the first fuse to be certified under 

the latest revision of the UL913 Safety Standard for Intrinsically 

Safe Apparatus and Associated Apparatus for Use 

in Class I, II, and III, Division 1, Hazardous 

(Classified) Locations. The PICO 259- 

UL913 Series is a range of encapsulated, 

hermetically sealed fuses that are 

ideal for 125 V applications in the oil, 

gas, mining, chemical, pharmaceutical, and 

food/beverage processing industries. 

The use of devices that are certified to 

be Intrinsically Safe (IS) is a requirement 

of operating electronic equipment in potentially 

explosive areas. The PICO 259-UL913 

Series Fuse is suitable for use in testing, measuring 

or processing electronic and electrical equipment, motor 

controllers, lighting, communication handsets, flow meters, process 

control and automation, sensors and other intrinsically safe 

Holy Stone 

A POWERFUL PRESENCE 

in Power Applications 

High Capacitance X7R for 

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High Voltage Capacitors for 

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High Capacitance MLCCs 

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Certified for Chargers 

and Adapters 

Stacked, High C/V for Switch 

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Holy Stone MLCCs are available with 

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Holy Stone Enterprise Co., Ltd. 

Tel: 951-696-4300 Fax: 951-696-4301 

info@holystonecaps.com www.holystonecaps.com 

In Europe: www.holystoneeurope.com In Asia: www.holystone.com.tw 

apparatus designed to operate in these hazardous locations. The 

PICO 259-UL913 Series Fuse and its encapsulant prevent heat 

and sparks from being exposed to the hazardous gases or dust in 

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apparatus for voltages not exceeding 125 Vrms (190 V peak). In 

addition to the Underwriters Laboratories, Inc. standard, these 

fuses meet ATEx, IECEx, and Baseefa requirements. 

Features 

• Hermetically sealed, encapsulated design 

• Well suited for 125 V applications 

• Current rating options from 62 mA to 5 A 

• Designed for operation in a range of hazardous environments 

• RoHS compliant 

Littelfuse PICO 259-UL913 Series Intrinsically Safe Fuses are 

packaged in bulk in quantities of 10 or 1,000 pieces. Samples 

are available now for bulk orders. 

Littelfuse Inc. 

Chicago, IL 

http://www.littelfuse.com. 

■Compact DC/DC Converters 

AVAILABLE IMMEDIATELY from MicroPower Direct, the 

G200EI series is a family of compact 2W, EN 60950 approved 

DC/DC converters. These converters are specifically designed 

for board level power applications that require small size, robust 

operation, high input/output isolation levels and low cost. 

Twenty one standard models operate from inputs of 5, 12 

or 24 VDC, providing single and dual outputs of 5, 9, 12, 

15, ±5, ±12, or ±15 VDC. 

Standard features include 

an input/output isolation of 

3,000 VDC, efficiency as 

high as 85%, and low noise 

operation. The MTBF (per 

MIL HDBK 217F) of the 

G200EI series is greater 

than 3.5 Mhours. All models 

are approved to EN 60950 and are RoHS compliant. 

The G200EI family is packaged in a miniature, Mini-DIP 

case that is only 0.787 x 0.394 x 0.4 in. The pin-out is industry 

standard and all case materials meet UL94-VO. Each model 

is specified for operation over the wide operating industrial 

temperature range of -40 °C to +85 °C with no derating or heat 

sinking required. The G200EI series is an ideal solution for a 

variety of board level power applications requiring small size, 

robust performance, and high reliability. 

MicroPower Direct 

Stoughton, MA 

http://www.micropowerdirect.com 


■Mercury-Free Nickel Zinc 

Batteries Meet OEM Needs 

ANNOUNCED BY VARTA Microbattery 

is the new mercury-free primary Nickel 

Zinc (NiOOH) Battery Systems. VARTA 

Microbattery’s nickel zinc cells are developed 

as an ideal cost effective and highly 

reliable substitute to conventional primary 

silver oxide (Ag2O) button cells. The cells 

are manufactured with high quality raw 

materials, unparalleled leakage protection, 

size, shape, and quality standards equivalent 

to VARTA Microbattery’s popular mercury-free 

primary SR Silver Oxide (Ag2O) 

System. As nickel zinc cell pricing is not 

tied to the volatility inherent in the silver 

market, end product pricing for retailers and 

OEMs is not adversely effected by material 

cost fluctuations. 

VARTA Microbattery’s nickel zinc cells 

may be specified for use in a wide range 

of miniature, low voltage 

power source applications 

in which highreliability,ecofriendlymercury-free 

cells 

are required. 

The ZR Nickel Zinc (NiOOH) 

System is particularly suited for watches 

(analog and digital) as it notably complies 

with the requirements of the international 

IEC 60086-3 standard for watch batteries. 

Moreover, the cells are highly appropriate 

for use in such medical devices as blood 

sugar indicators, insulin pumps and thermometers, 

as well as in electronic calculators, 

laser pointers, toys, and remote control 

applications. 

The nickel zinc cells feature a voltage level 

of 1.65V and are available with 8 mAh to 

130 mAh capacity, in heights ranging from 

2.10mm to 5.4mm, and in weights from 

0.23 g to 2.33 g. The cells additionally 

boast a stable discharge curve, good pulse 

load behavior, and an EOL detection feature 

for watch applications. For more detailed 

information regarding VARTA Microbattery’s 

cells and configurations, visit http://www. 

varta-microbattery.com/en/products/batteries-cells-configurations.html. 

Pricing for VARTA Microbattery’s mercury-free 

primary ZR Nickel Zinc (NiOOH) 

Button 364 Cell starts at $0.13 each in 

quantities of 10,000. Delivery time is to 

2-4 weeks. 

VARTA Microbattery 

White Plains, NY 

http://www.varta-microbattery.com 


POWERelectronics 

SAM DAVIS, Senior Technical Editor 

POWER BRIDGE CIRCUIT FOR BI-DIRECTIONAL 

INDUCTIVE SIGNALING 

Issued: May 7, 2013 

United States Patent 8,437,695 

An inductive signal interface comprises a coil assembly including one or 

more inductive coils, a bridge circuit including a plurality of switches, 

and control circuitry. The control circuitry is configured to individually 

operate the plurality of switches to enable the inductive signal interface 

to dynamically switch between a power-transmit mode and a power 

receive mode. 

Inventors: Chatterjee; Manjirnath (San Francisco, CA), Lehr; Michael 

(Sunnyvale, CA), Shaffer; Dyke (Santa Rosa, CA) 

Assignee: Hewlett-Packard Development Company, L.P. (Houston, TX) 

Appl. No.: 12/841,001 

Filed: July 21, 2010 

WIDE INPUT VOLTAGE RANGE POWER FACTOR 

CORRECTION CIRCUIT 



A boost circuit is used for power factor correction (PFC). In a low power 

application, transition mode control is utilized. However, switching frequency 

varies with different input voltages, and over a wide input voltage 

range, the switching frequency can become too high to be practical. To 

address this issue, a boost circuit is provided whose effective inductance 

changes as a function of input voltage. By changing the inductance, control 

is exercised over switching frequency. 

Inventors: Shao; Jianwen (Hoffman Estates, IL), Hopkins; Thomas Lea 

R. (Mundelein, IL) 

Assignee: STMicroelectronics, Inc. (Coppell, TX) 

Appl. No.: 13/023,072 

Filed: February 8, 2011 

BATTERY CELL EQUALIZER SYSTEM 



A method of operating a battery system includes a plurality of battery 

cells coupled in series. The plurality of cells includes at least three battery 

cells coupled in series. The method includes determining a cell with the 

greatest charge excess of the plurality of battery cells. The method further 

includes determining a cell with the greatest charge deficit of the plurality 

of battery cells. The method further includes discharging the cell with the 

greatest charge excess to charge, with a voltage converter, the cell with the 

greatest charge deficit. 

Inventors: Pigott; John M. (Phoenix, AZ) 

Assignee: Freescale Semiconductor, Inc. (Austin, TX) 

Appl. No.: 12/833,430 


SYSTEMS AND METHODS FOR IMPROVED 

OVER-CURRENT CLIPPING 



Systems and methods for implementing over-current protection include 

reducing a clip level while an over-current condition is being detected. 

Once the over-current condition is no longer detected, the clip level is 

maintained for a specified period before allowing the clip level to be 

increased. In an embodiment, the specified period, for which the clip level 

is maintained before the clip level is allowed to be increased, starts when 

the over-current condition is no longer detected, and ends when each of 

N immediately preceding sample(s) of the audio signal are not clipped to 

the clip level, where N is an integer. After an over-current condition is no 

longer detected, and after the clip level has been maintained for the specified 

period, the clip level can be increased if an over-current condition is 

not detected for a sample and the clip level is below a specified maximum 

clip level. 

Inventors: Kost; Michael A. (Cedar Park, TX) 

Assignee: Intersil Americas Inc. (Milpitas, CA) 

Appl. No.: 13/284,723 

Filed: October 28, 2011 

LEVEL CONVERTER CIRCUIT FOR USE IN CMOS CIRCUIT 

DEVICE PROVIDED FOR CONVERTING SIGNAL LEVEL OF 

DIGITAL SIGNAL TO HIGHER LEVEL 



A level converter circuit is provided for converting an input signal of 

a digital signal having a first signal level into an output signal having a 

second signal level higher than the first signal level. An amplifier circuit 

amplifies the input signal and outputs an amplified output signal, and 

a current generator circuit generates a control current corresponding to 

an operating current flowing through the amplifier circuit upon change 

of the signal level of the input signal. A current detector circuit detects 

the generated control current, and controls the operating current of the 

amplifier circuit to correspond to the detected control current. The current 

generator circuit includes series-connected first and second nMOS 

transistors as inserted between the current detector circuit and the ground. 

The first nMOS transistor operates responsive to the input signal, and the 

second nMOS transistor operates responsive to an inverted signal of the 

input signal. 

Inventors: Hirose; Tetsuya (Kobe, JP), Osaki; Yuji (Kobe, JP), Mori; 

Toshihiko (Yokohama, JP) 

Assignee: Semiconductor Technology Academic Research Center 

(Kanagawa, JP) 

Appl. No.: 13/181,825 


APPARATUS FOR DETECTING A STATE OF OPERATION 

OF A POWER SEMICONDUCTOR DEVICE 



An embodiment of the invention relates to an apparatus including a power 

semiconductor device and a processor coupled thereto. The processor is 

configured to provide a control signal to the power semiconductor device 

to regulate an output characteristic of the apparatus. The processor models 

an internal characteristic of the power semiconductor device and alters 

the control signal if the modeled internal characteristic crosses a threshold 

value. In an exemplary embodiment, the internal characteristic is a channel 

temperature of a MOSFET. A sensor such as a thermistor is coupled 

to or included within the processor to sense a parameter separate from 

the power semiconductor device, such as a processor temperature, and 

the processor is configured to adapt the modeled internal characteristic to 

the sensed parameter. 

Inventors: Pelz; Georg (Ebersberg, DE), Lenz; Michael (Zorneding, DE), 

Kunze; Matthias (Neubiberg, DE) 

Assignee: Infineon Technologies Austria AG (Villach, AT) 

Appl. No.: 13/088,200 

Filed: April 15, 2011 


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