The superhet or superheterodyne radio receiver

The superhet or superheterodyne radio receiver
- an introduction to the operation of the superheterodyne radio receiver
and how it uses the process of mixing and frequency translation with an
intermediate frequency amplifier and filter to provide high levels of
selectivity and amplification
The superhet radio or to give it its full name the superheterodyne receiver is one of the
most popular forms of receiver in use today. Virtually all broadcast radios, televisions
and many more types of receiver use the superhet or superheterodyne principle. First
developed at the end of the First World War, with its invention credited to the American
Edwin Armstrong, the use of the superhet has grown ever since the concept was first
discovered.
Mixing
The idea of the superhet revolves around the process of mixing. Here RF mixers are used
to multiply two signals together. (This is not the same as mixers used in audio desks
where the signals are added together). When two signals are multiplied together the
output is the product of the instantaneous level of the signal at one input and the
instantaneous level of the signal at the other input. It is found that the output contains
signals at frequencies other than the two input frequencies. New signals are seen at
frequencies that are the sum and difference of the two input signals, i.e. if the two input
frequencies are f1 and f2, then new signals are seen at frequencies of (f1+f2) and (f1-f2).
To take an example, if two signals, one at a frequency of 5 MHz and another at a
frequency of 6 MHz are mixed together then new signals at frequencies of 11 MHz and 1
MHz are generated.
The signals generated by mixing or multiplying two signals together
Concept of the superheterodyne receiver
In the superhet or superheterodyne radio, the received signal enters one input of the
mixed. A locally generated signal (local oscillator signal) is fed into the other. The result
is that new signals are generated. These are applied to a fixed frequency intermediate
frequency (IF) amplifier and filter. Any signals that are converted down and then fall
within the passband of the IF amplifier will be amplified and passed on to the next stages.
Those that fall outside the passband of the IF are rejected. Tuning is accomplished very
simply by varying the frequency of the local oscillator. The advantage of this process is
that very selective fixed frequency filters can be used and these far out perform any
variable frequency ones. They are also normally at a lower frequency than the incoming
signal and again this enables their performance to be better and less costly.
To see how this operates in reality take the example of two signals, one at 6 MHz and
another at 6.1 MHz. Also take the example of an IF situated at 1 MHz. If the local
oscillator is set to 5 MHz, then the two signals generated by the mixer as a result of the 6
MHz signal fall at 1 MHz and 11 MHz. Naturally the 11 MHz signal is rejected, but the
one at 1 MHz passes through the IF stages. The signal at 6.1 MHz produces a signal at
1.1 MHz (and 11.1 MHz) and this falls outside bandwidth of the IF so the only signal to
pass through the IF is that from the signal on 6 MHz.

The basic concept of the superhet radio
If the local oscillator frequency is moved up by 0.1 MHz to 5.1 MHz then the signal at
6.1 MHz will give rise to a signal at 1 MHz and this will pass through the IF. The signal
at 6 MHz will give rise to a signal of 0.9 MHz at the IF and will be rejected. In this way
the receiver acts as a variable frequency filter, and tuning is accomplished.
Images
The basic concept of the superheterodyne receiver appears to be fine, but there is a
problem. There are two signals that can enter the IF. With the local oscillator set to 5
MHz and with an IF it has already been seen that a signal at 6 MHz mixes with the local
oscillator to produce a signal at 1 MHz that will pass through the IF filter. However if a
signal at 4 MHz enters the mixer it produces two mix products, namely one at the sum
frequency which is 10 MHz, whilst the difference frequency appears at 1 MHz. This
would prove to be a problem because it is perfectly possible for two signals on
completely different frequencies to enter the IF. The unwanted frequency is known as the
image. Fortunately it is possible to place a tuned circuit before the mixer to prevent the
signal entering the mixer, or more correctly reduce its level to acceptable value.
Fortunately this tuned circuit does not need to be very sharp. It does not need to reject
signals on adjacent channels, but instead it needs to reject signals on the image frequency.
These will be separated from the wanted channel by a frequency equal to twice the IF. In
other words with an IG at 1 MHz, the image will be 2 MHz away from the wanted
frequency.
Using a tuned circuit to remove the image signal
Complete receiver
Having looked at the concepts behind the superheterodyne receiver it is helpful to look at
a block diagram of a basic superhet. Signals enter the front end circuitry from the
antenna. This contains the front end tuning for the superhet to remove the image signal
and often includes an RF amplifier to amplify the signals before they enter the mixer. The
level of this amplification is carefully calculated so that it does not overload the mixer

when strong signals are present, but enables the signals to be amplified sufficiently to
ensure a good signal to noise ratio is achieved.
The tuned and amplified signal then enters one port of the mixer. The local oscillator
signal enters the other port. The local oscillator may consist of a variable frequency
oscillator that can be tuned by altering the setting on a variable capacitor. Alternatively it
may be a frequency synthesizer that will enable greater levels of stability and setting
accuracy.
Once the signals leave the mixer they enter the IF stages. These stages contain most of
the amplification in the receiver as well as the filtering that enables signals on one
frequency to be separated from those on the next. Filters may consist simply of LC tuned
transformers providing inter-stage coupling, or they may be much higher performance
ceramic or even crystal filters, dependent upon what is required.
Once the signals have passed through the IF stages of the superheterodyne receiver, they
need to be demodulated. Different demodulators are required for different types of
transmission, and as a result some receivers may have a variety of demodulators that can
be switched in to accommodate the different types of transmission that are to be
encountered. The output from the demodulator is the recovered audio. This is passed into
the audio stages where they are amplified and presented to the headphones or
loudspeaker.
Block diagram of a basic superheterodyne receiver
The diagram above shows a very basic version of the superhet or superheterodyne
receiver. Many sets these days are far more complicated. Some superhet radios have
more than one frequency conversion, and other areas of additional circuitry to provide the
required levels of performance. However the basic superheterodyne concept remains the
same, using the idea of mixing the incoming signal with a locally generated oscillation to
convert the signals to a new frequency.
Selectivity is one of the major specifications of any receiver. Whilst the sensitivity is
important to ensure that it can pick up the signals and receive them at a sufficient
strength, the selectivity is also very important. It is this parameter that determines
whether the receiver is able to pick out the wanted signal from all the other ones around
it. The filters used in receivers these days have very high levels of performance and
enable receivers to select out individual signals even on today's crowded bands.
Superhet principle
Most of the receivers that are used today are superhet radios. In these sets the incoming
signal is converted down to a fixed intermediate frequency. It is within the IF stages that
the main filters are to be found. It is the filter in the IF stages that defines the selectivity
performance of the whole set, and as a result the receiver selectivity specification is
virtually that of the filter itself.

Block diagram of a basic superhet receiver
In some receivers simple LC filters may be used, although ceramic filters are better and
are used more widely nowadays. For the highest performance crystal or mechanical
filters may be used, although they are naturally more costly and this means they are only
found in high performance sets.
Filter parameters
There are two main areas of interest for a filter, the pass band where it accepts signals and
allows them through, and the stop band where it rejects them. In an ideal world a filter
would have a response something like that shown in Figure 2. Here it can be seen that
there is an immediate transition between the pass band and the stop band. Also in the pass
band the filter does not introduce any loss and in the stop band no signal is allowed
through.
The response of an ideal filter
In reality it is not possible to realise a filter with these characteristics and a typical
response more like that shown in Figure 3. It is fairly obvious from the diagram that there
are a number of differences. The first is that there is some loss in the pass band. Secondly
the response does not fall away infinitely fast. Thirdly the stop band attenuation is not
infinite, even though it is very large. Finally it will be noticed that there is some in band
ripple.
Typical response of a real filter

In most filters the attenuation in the pass band is normally relatively small. For a typical
crystal filter figures of 2 - 3 dB are fairly typical. However it is found that very narrow
band filters like those used for Morse reception may be higher than this. Fortunately it is
quite easy to counteract this loss simply by adding a little extra amplification in the
intermediate frequency stages and this factor is not quoted as part of the receiver
specification.
It can be seen that the filter response does not fall away infinitely fast, and it is necessary
to define the points between which the pass band lies. For receivers the pass band is taken
to be the bandwidth between the points where the response has fallen by 6 dB, i.e. where
it is 6 dB down or -6 dB.
A stop band is also defined. For most receiver filters this is taken to start at the point
where the response has fallen by 60 dB, although the specification for the filter should be
checked this as some filters may not be as good. Sometimes a filter may have the stop
band defined for a 50 dB attenuation rather than 60 dB.
Shape factor
It can be seen that it is very important for the filter to achieve its final level of rejection as
quickly as possible once outside the pass band. In other words the response should fall as
quickly as possible. To put a measure on this, a figure known as the shape factor is used.
This is simply a ratio of the bandwidths of the pass band and the stop band. Thus a filter
with a pass band of 3 kHz at -6dB and a figure of 6 kHz at -60 dB for the stop band
would have a shape factor of 2:1. For this figure to have real meaning the two attenuation
figures should also be quoted. As a result the full shape factor specification should be 2:1
at 6/60 dB.
Filter types
There is a variety of different types of filter that can be used in a receiver. The older
broadcast sets used LC filters. The IF transformers in the receiver were tuned and it was
possible to adjust the resonant frequency of each transformer using an adjustable ferrite
core.
Today ceramic filters are more widely used. Their operation is based on the piezoelectric
effect. The incoming electrical signal is converted into mechanical vibrations by the
piezoelectric effect. These vibrations are then affected by the mechanical resonances of
the ceramic crystal. As the mechanical vibrations are then linked back to the electric
signal, the overall effect is that the mechanical resonances of the ceramic crystal affect
the electrical signal. The mechanical resonances of the ceramic exhibit a high level of Q
and this is reflected in its performance as an electrical filter. In this way a high Q filter
can be manufactured very easily.
Ceramic filters can be very cheap, some costing only a few cents. However higher
performance ones are also available, and these are likely to be found in scanners and
many other receivers.
For really high levels of filter performance crystal filters are used. Crystals are made
from quartz, a naturally occurring form of silicon, although today's components are made
from synthetically grown quartz. These crystals also use the piezoelectric effect and
operate in the same way as ceramic filters but they exhibit much higher levels of Q and
offer far superior degrees of selectivity. Being a resonant element they are used in many
areas where an LC resonant element might be found. They are used in oscillators - many
computers have crystal oscillators in them, but they are also widely used in high
performance filters.
Normally crystal filters are made from a number of individual crystals. The most
commonly used configuration is called the half lattice filter as shown in Figure 4. Further
sections can be added to the filter to improve the performance. Often a filter will be
quoted as having a certain number of poles. There is one pole per crystal, so a six pole
crystal filter would contain six crystals and so forth. Many filters used in amateur
communications receivers will contain either six or eight poles.

A basic half lattice crystal filter section
Choosing the right bandwidth
It is important to choose the correct bandwidth for a give type of signal. It is obviously
necessary to ensure that it is not too wide, otherwise unwanted off-channel signals will be
able to pass though the filter. Conversely if the filter is too narrow then some of the
wanted signal will be rejected and distortion will occur. As different types of
transmission occupy different amounts of spectrum bandwidth it is necessary to tailor the
filter bandwidth to the type of transmission being received. As a result many receivers
switch in different filters for different types of transmission. This may be done either
automatically as part of a mode switch, or using a separate filter switch. Typically a filter
for AM reception on the short wave bands will have a bandwidth of around 6 kHz, and
one for SSB will be approximately 2.5 kHz. For Morse reception 500 and 250 Hz filters
are often used.
Summary
Selectivity is particularly important on today's crowded bands, and it is necessary to
ensure that any receiver is able to select the wanted signal as well as it can. Obviously
when signals occupy the same frequency there is little that can be done, but by having a
good filter it is possible to ensure that you have the best chance or receiving and being
able to copy the signal you want.
he superhet radio receiver is one of the most widely used types of receiver available. One
of the important specifications associated with its operation is image response or image
rejection. Along with this the IF breakthrough is also of importance, although less critical
in many applications.
Image response
The basic concept of the superhet radio means that it is possible for two signals to eneter
the intermediate frequency (IF) implifier. For example with the local oscillator set to 5
MHz and with an IF of 1 MHz it can be seen that a signal at 6 MHz mixes with the local
oscillator to produce a signal at 1 MHz that will pass through the IF filter. However is a
signal at 4 MHz is also able to produce an output at 1 MHz. It is clearly unacceptable to
receive signals on two frequencies at the same time and it is possible to remove the
unwanted one by the addition of a tuned circuit prior to the mixer
Fortunately this tuned circuit does not need to be excessively sharp. It does not need to
reject signals on adjacent channels, but instead it needs to reject signals on the image
frequency. These will be separated from the wanted channel by a frequency equal to
twice the IF. In other words with an IG at 1 MHz, the image will be 2 MHz away from
the wanted frequency.

Using a tuned circuit to remove the image signal
Image
It is clearly important to specify the level of rejection of the image signal. The
specification compares the levels of signals of equal strength on the wanted and image
frequencies, quoting the level of rejection of the unwanted signal.
The image rejection of a receiver will be specified as the ratio between the wanted and
image signals expressed in decibels (dB)at a certain operating frequency. For example it
may be 60 dB at 30 MHz. This means that if signals of the same strength were present on
the wanted frequency and the image frequency, then the image signal would be 60 dB
lower than the wanted one, i.e. it would be 1/1000 lower in terms of voltage or 1/1000000
lower in terms of power.
The frequency at which the measurement is made also has to be included. This is because
the level of rejection will vary according to the frequency in use. Typically it falls with
increasing frequency because the percentage frequency difference between the wanted
and image signals is smaller.
IF Breakthrough
Another problem which can occur with a superhet occurs when signals from the aerial
break through the RF sections of the set and directly enter the IF stages. Normally
intermediate frequencies are chosen so that there are likely to be no very large signals
present which might cause problems. However when the receiver has a fixed frequency
first local oscillator this is not easy to ensure as it will sweep over a band of frequencies.
The specification for breakthrough is quoted in the same fashion as image rejection.
Normally it is possible to achieve figures of 60 to 80 dB rejection, and on some receivers
figures of 100 dB have been quoted.
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Radio receiver sensitivity
- including the concept of noise and sensitivity, signal to noise ratio,
SINAD, and noise figure.
Receiver sensitivity is one of the key specifications of any radio. The two main
requirements of any radio receiver are that it should be able to separate one station from
another, i.e. selectivity, and signals should be amplified so that they can be brought to a
sufficient level to be heard. As a result receiver designers battle with many elements to
make sure that these requirements are fulfilled
A number of methods of measuring and specifying the sensitivity performance of radio
receivers are used. Figures including signal to noise ratio, SINAD, noise factor and noise
figure are used. These all use the fact that the limiting factor of the sensitivity of a radio
receiver is not the level of amplification available, but the levels of noise that are present,
whether they are generated within the radio receiver or outside it.

Noise
Today technology is such that there is little problem in being able to achieve very large
levels of amplification within a radio receiver. This is not the limiting factor. In any
receiving station the limiting factor is noise - weak signals are not limited by the actual
signal level, but by the noise masks them out. This noise can come from a variety of
sources. It can be picked up by the antenna or it can be generated within the radio
receiver.
It is found that the level of noise that is picked up externally by a receiver from the
antenna falls as the frequency increases. At HF and frequencies below this the
combination of galactic, atmospheric and man-made noise is relatively high and this
means that there is little point in making a receiver particularly sensitive. Normally radio
receivers are designed such that the internally generated noise is much lower than any
received noise, even for the quietest locations.
At frequencies above 30 MHz the levels of noise start to reach a point where the noise
generated within the radio receiver becomes far more important. By improving the noise
performance of the radio receiver, it becomes possible to detect much weaker signals.
Design for noise performance
In terms of the receiver noise performance it is always the first stages or front end that is
most crucial. At the front end the signal levels are at their lowest and even very small
amounts of noise can be comparable with the incoming signal. At later stages in the set
the signal will have been amplified and will be much larger. The same levels of noise as
are present at the front end will be a much smaller proportion of the signal and will not
have the same effect. Accordingly it is important that the noise performance of the front
end is optimised for its noise performance.
It is for this reason that the noise performance of the first radio frequency amplifier
within the receiver is of great importance. It is the performance of this circuit that is
crucial in determining the performance of the whole radio receiver. To achieve the
optimum performance for the first stage of the radio receiver there are a number of steps
that can be taken. These include:
• Determine the circuit topology required
• Choose a low noise device
• Determine the gain required
• Determine the current through the device
• Use low noise resistors
• Optimise the matching
• Ensure that power supply noise entering the circuit is removed
Determination of circuit topology The first step in any design is to decide upon the type
of circuit to be used. Whether a conventional common emitter style circuit is to be used,
or even whether a common base should be employed. The decision will depend upon
factors including the matching input and output impedances, the level of gain required
and the matching arrangements to be used.
Choice of active device The type of device to be used is also important. There are
generally two decisions, whether to use a bipolar based transistor, or whether to use a
field effect device. Having made this, it is obviously necessary to decide upon a low
noise device. The noise performance of transistors and FETs is normally specified, and
special high performance low noise devices are available for these applications.
Determination of required gain While it may appear that the maximum level of gain
may be required from this stage to minimise the levels of amplification required later and
in this way ensure that the noise performance is optimised, this is not always the case.
There are two major reasons for this. The first is that the noise performance of the circuit
may be impaired by requiring too high a level of gain. Secondly it may lead to overload
in later stages of the radio receiver and this may degrade the overall performance. Thus
the level of gain required must be determined from the fact that it is necessary to optimise

the noise performance of this stage, and secondly to ensure that later stages of the
receiver are not overloaded.
Determination of current through the active device The design of the first stage of the
radio receiver must be undertaken with care. To obtain the required RF performance in
terms of bandwidth and gain, it may be necessary to run the device with a relatively high
level of current. This will not always be conducive to obtaining the optimum noise
performance. Accordingly the design must be carefully optimised to ensure the best
performance for the whole radio receiver.
Use of low noise resistors It may appear to be an obvious statement, but apart from
choosing a low noise active device, consideration should also be given to the other
components in the circuit. The other chief contributors are the resistors. The metal oxide
film resistors used these days, including most surface mount resistors normally offer
good performance in this respect and can be used as required.
Optimise impedance matching In order to obtain the best noise performance for the
whole radio receiver it is necessary to optimise the impedance matching. It may be
thought that it is necessary to obtain a perfect impedance match. Unfortunately the best
noise performance does not usually coincide with the optimum impedance match
Accordingly during the design of the RF amplifier it is necessary to undertake some
design optimisation to ensure the best overall performance is achieved for the radio
receiver.
Ensure that power supply noise entering the circuit is removed Power supplies can
generate noise. In view of this it is necessary to ensure that any noise generated by the
radio receiver power supply does not enter the RF stage. This can be achieved by
ensuring that there is adequate filtering on the supply line to the RF amplifier.
Summary
Receiver sensitivity is one of the vital specifications of any radio receiver. The key factor
in determining the sensitivity performance of the whole receiver is the RF amplifier. By
optimising its performance, the figures for the whole of the receiver can be improved. In
this way the specifications for signal to noise ratio, SINAD or noise figure can be brought
to the required level.
There are a number of ways in which the noise performance, and hence the sensitivity of
a radio receiver can be measured. The most obvious method is to compare the signal and
noise levels for a known signal level, i.e. the signal to noise (S/N) ratio or SNR.
Obviously the greater the difference between the signal and the unwanted noise, i.e. the
greater the S/N ratio, the better the radio receiver sensitivity performance.
As with any sensitivity measurement, the performance of the overall radio receiver is
determined by the performance of the front end RF amplifier stage. Any noise introduced
by the first RF amplifier will be added to the signal and amplified by subsequent
amplifiers in the receiver. As the noise introduced by the first RF amplifier will be
amplified the most, this RF amplifier becomes the most critical in terms of radio receiver
sensitivity performance. Thus the first amplifier of any radio receiver should be a low
noise amplifier.
Methods of measuring receiver sensitivity
Although there are many ways of measuring the sensitivity performance of a radio
receiver, the S/N ratio or SNR is one of the most straightforward and it is used in a
variety of applications. However it has a number of limitations, and although it is widely
used, other methods including noise figure are often used as well. Nevertheless the S/N
ratio or SNR is an important specification, and it will be seen in many radio receiver
specification sheets.

Signal to noise ratio for a radio receiver
The difference is normally shown as a ratio between the signal and the noise (S/N) and it
is normally expressed in decibels. As the signal input level obviously has an effect on this
ratio, the input signal level must be given. This is usually expressed in microvolts.
Typically a certain input level required to give a 10 dB signal to noise ratio is specified.
Effect of bandwidth
A number of other factors apart from the basic performance of the set can affect the SNR
specification. The first is the actual bandwidth of the receiver. As the noise spreads out
over all frequencies it is found that the wider the bandwidth of the receiver, the greater
the level of the noise. Accordingly the receiver bandwidth needs to be stated.
Additionally it is found that when using AM the level of modulation has an effect. The
greater the level of modulation, the higher the audio output from the receiver. When
measuring the noise performance the audio output from the receiver is measured and
accordingly the modulation level of the AM has an effect. Usually a modulation level of
30% is chosen for this measurement.
Typical figures
This method of measuring the performance is most commonly used for HF
communications receivers. Typically one might expect to see a figure in the region of 0.5
microvolts for a 10 dB S/N in a 3 kHz bandwidth for SSB or Morse. For AM a figure of
1.5 microvolts for a 10 dB S/N in a 6 kHz bandwidth at 30% modulation for AM might
be seen.
Points to note when measuring SNR
SNR is a very convenient method of quantifying the sensitivity of a receiver, but there are
some points to note when measuring and interpreting the figures. To investigate these it is
necessary to look at the way the measurements of SNR are made. A calibrated RF signal
generator is used as a signal source for the receiver. It must have an accurate method of
setting the output level down to very low signal levels. Then at the output of the receiver
a true RMS AC voltmeter is used to measure the output level.
S/N and (S+N)/N With the generator signal switched off a 50 Ohm match is
given to the receiver and the audio meter will detect the noise generated by the
receiver itself. This level is noted and the signal turned on. Its level is adjusted
until the audio level meter reads a level which is 10 dB higher than just the noise
on its own. The level of the generator is that required to give the 10 dB signal to
noise ratio.
The last statement was not strictly true. Whilst the first reading of the noise is
quite accurate, the second reading of the signal also includes some noise as well.
In view of this many manufacturers will specify a slightly different ratio: namely
signal plus noise to noise (S+N/N). In practice the difference is not particularly
large, but the S+N/N ratio is more correct.
PD and EMF Occasionally the signal generator level in the specification will
mention that it is either PD or EMF. This is actually very important because there
is a factor of 2:1 between the two levels. For example 1 microvolt EMF. and 0.5
microvolt PD are the same. The EMF (electro-motive force) is the open circuit

voltage, whereas the PD (potential difference) is measured when the generator is
loaded. As a result of the way in which the generator level circuitry works it
assumes that a correct (50 Ohm) load has been applied. If the load is not this
value then there will be an error. Despite this most equipment will assume values
in PD unless otherwise stated.
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Radio receiver SINAD measurement
- an overview of the SINAD measurement used in specifying the sensitivity
performance of many radio receivers.
One of the measurements that can be made to assess and specify the sensitivity
performance of a radio receiver is SINAD. While not used as widely as the signal to
noise ratio, or noise figure it is nevertheless used commonly and can be found in the
specifications of many radio receivers. SINAD is often used in conjunction with FM
receivers, but it can also be used for AM and SSB quite easily.
As with any radio receiver, the design of the RF amplifier is key to its sensitivity
performance. A poorly performing RF amplifier will degrade the performance of the
whole receiver. However a high performance low noise RF amplifier will enable the
overall set to provide a high level of sensitivity. Accordingly time should be focussed in
the design of the RF amplifier in order that it should reach the required level of
performance.
What is SINAD?
SINAD is a measurement that can be used for any communication device to look at the
degradation of the signal by unwanted or extraneous signals including noise and
distortion. However the SINAD measurement is most widely used for measuring and
specifying the sensitivity of a radio receiver.
The actual definition of SINAD is quite straightforward. It can be summarised as the ratio
of the total signal power level (Signal + Noise + Distortion) to unwanted signal power
(Noise + Distortion). Accordingly, the higher the figure for SINAD, the better the quality
of the audio signal.
The SINAD figure is expressed in decibels (dB) and can be determined from the simple
formula:
SINAD = 10Log ( SND / ND )
where:
SND = combined Signal + Noise + Distortion power level
ND = combined Noise + Distortion power level
It is worth noting that SINAD is a power ratio and not a voltage ratio for this calculation.
Making SINAD measurements
To make the measurement a signal modulated with an audio tone is entered into the radio
receiver. A frequency of 1 kHz is taken as the standard as it falls in the middle of the
audio bandwidth. A measurement of the whole signal, i.e. the signal plus noise plus
distortion is made. As the frequency of the tone is known, the regenerated audio is passed
through a notch filter to remove the tone. The remaining noise and distortion is then
measured.
Although it is most common to measure the electrical output at the receiver audio output
terminals, another approach that is not as widely used, is to pass the signal into the
loudspeaker and then use a transducer connected to SINAD meter to convert the audio
back into an electrical signal. This will ensure that any distortion included by the speaker

is incorporated, and it may overcome problems with gaining access to the speaker
connections in certain circumstances where this may not be possible.
Obtaining the figures for the signal plus noise plus distortion and the noise plus distortion
it is then possible to calculate the value of SINAD for the radio receiver of other piece of
equipment.
The set up used for making SINAD measurements
While the measurements for SINAD can be made using individual items of test
equipment, a number of SINAD meters are made commercially. These SINAD meters
incorporate all the required circuitry and can be connected directly to radio receivers to
make the measurements. Accordingly SINAD meters are a particularly convenient
method of making these measurements.
Filter for SINAD measurements
The notch filter that is required for SINAD measurements to be made has an effect on the
measurement. In an ideal world the filter would be infinitely sharp a notch out only the
modulating tone. However in the real world the filter will have a finite bandwidth. As its
bandwidth increases, so it will remove noise and distortion as well as the tone. However
as the distortion products will typically result from the second and third harmonics of the
tone, the filter will not have an effect on this element of the reading. Nevertheless it may
still have an effect on the noise levels.
In view of this problem some standards set down specifications or guidelines for the filter
used in the SINAD measurement. ETSI (European Telecommunications Standards
Institute) defines a notch filter (ETR 027). With the standard tone frequency of 1 kHz, it
states that a filter used for SINAD measurements shall be such that the output the 1000
Hz tone shall be attenuated by at least 40 dB and at 2000 Hz the attenuation shall not
exceed 0.6 dB. The filter characteristic shall be flat within 0.6 dB over the ranges 20 Hz
to 500 Hz and 2000 Hz to 4000 Hz. In the absence of modulation the filter shall not cause
more than 1 dB attenuation of the total noise power of the audio frequency output of the
receiver under test.
In addition to the filter performance another critical area of a SINAD measurement is the
measurement of the output signal power levels. These have to be a true power
measurements that accommodate the different form factors of the variety of waveforms,
i.e. sine wave for the 1 kHz tone and its harmonics, but the noise will be random and
have a different form factor.
Applications of SINAD measurements
SINAD measurements give an assessment of the signal quality from a receiver under a
number of conditions. As such SINAD measurements can be used for assessing a number
of elements of receiver performance.
Receiver sensitivity: The most common use of the SINAD measurement is to assess the
sensitivity performance of a radio receiver. To achieve this the sensitivity can be assessed
by determining the RF input level at the antenna that is required to achieve a given figure
of SINAD. Normally a SINAD value of 12 dB is taken as this corresponds to a distortion
factor of 25%, and a modulating tone of 1 kHz is used. It is also necessary to determine
other conditions. For AM it is necessary to specify the depth of modulation and for FM

the level of deviation is required. For FM analogue systems ETSI specifies the use of a
deviation level of 12.5% of the channel spacing
A typical specification might be that a receiver has a sensitivity of 0.25 uV [microvolts]
for a 12 dB SINAD. Obviously the lower the input voltage needed to achieve the given
level of SINAD, the better the receiver performance.
Adjacent channel rejection: This parameter is a measure of the ability of the receiver to
reject signals on a nearby channel. As the adjacent channel performance degrades, so the
levels of noise and extraneous signals will increase, thereby degrading the SINAD
performance.
An initial measurement of SINAD is made at a given level and this is known as the
reference sensitivity. The RF input level of the signal for the SINAD measurement is then
increased by 3 dB at the receiver antenna input. A second source or signal with
modulated with a 400 Hz tone is added with its frequency set to an adjacent channel or at
a specific offset from the carrier source used for the basic SINAD measurement. It will be
found that the interferer will cause the 400 Hz tone to appear in the audio of the receiver
as its level is increased. This will be seen as a degradation in the SINAD as the 400 Hz
tone will pass through the SINAD meter notch filter.
With the measurement system set up, the interferer signal level is raised until the SINAD
value is degraded to the original value obtained at the reference sensitivity. Then the ratio
of the interfering level to the wanted signal is the adjacent channel rejection.
Receiver blocking: SINAD can be used to form the basis of a receiver blocking
measurement. As with other similar measurements a reference SINAD sensitivity level is
found. The level of the SINAD signal is increased by 3 dB at the antenna. An unmodulated
off channel signal is then added and its level raised until the receiver
desensitises to an extent whereby the reference SINAD level is reached.
Summary
SINAD is a particularly useful measurement format that can be used to determine the
performance of a radio receiver under a variety of conditions. Although SINAD is
primarily used to specify the basic sensitivity performance of many radios, it can be used
for other parameters as well. Additionally it is chiefly used for FM systems, but its use is
equally applicable to AM and SSB. It may also be used for digital systems as well,
although this is not common practice as a measurement known as bit error rate (BER) is
more widely used.
The overall figure for SINAD will be chiefly dependent upon the performance of the RF
amplifier in the receiver. A low noise RF amplifier will enable the set as a whole to
provide a good SINAD performance.
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Radio receiver noise figure
- an overview of noise figure used in specifying the sensitivity performance
of radio receivers and their components.
Although there are a number of methods of determining the sensitivity of radio receivers
and their associated elements, the noise figure is one of the most widely used methods.
Not only is it widely used to assess the sensitivity performance or receivers, but it can be
applied to complete receiving systems or to elements such as RF amplifiers. Thus it is
possible to use the same notation to measure the noise performance of a whole receiver,
or an RF amplifier. This makes it possible to determine whether a low noise amplifier
may be suitable for a particular system by judging their relative levels of performance.

Basics
Essentially the measurement assesses the amount of noise each part of the system or the
system as a whole introduces. This could be the radio receiver, or an RF amplifier for
example. If the system were perfect then no noise would be added to the signal when it
passed through the system and the signal to noise ratio would be the same at the output as
at the input. As we all know this is not the case and some noise is always added. This
means that the signal to noise ratio or SNR at the output is worse than the signal to noise
ratio at the input. In fact the noise figure is simply the comparison of the SNR at the input
and the output of the circuit.
A figure known as the noise factor can be derived simply by taking the SNR at the input
and dividing it by the SNR at the output. As the SNR at the output will always be worse,
i.e. lower, this means that the noise factor is always greater than one.
The noise factor is rarely seen in specifications. Instead the noise figure is always seen.
This is simply the noise factor expressed in decibels.
Noise figure
In the diagram S1 is the signal at the input, N1 is the noise at the input
and S2 is the signal at the output and N2 the noise at the output
As an example if the signal to noise ratio at the input was 4:1, and it was 3:1 at the output
then this would give a noise factor of 4/3 and a noise figure of 10 log (4/3) or 1.25 dB.
Alternatively if the signal to noise ratios are expressed in decibels then it is quite easy to
calculate the noise figure simply by subtracting one from another because two numbers
are divided by subtracting their logarithms. In other words if the signal to noise ratio was
13 dB at the input and only 11 dB at the output then the circuit would have a noise figure
of 13 - 11 or 2 dB.
Typical examples
The specifications of different pieces of equipment will vary quite widely. A typical HF
receiver may have a noise figure of 15 dB of more and function quite satisfactorily. A
better level of performance is not necessary because of the high level of atmospheric
noise. However an amateur receiver used on Two metres, for example, might have a
noise figure of 3 or 4 dB. RF amplifiers for this band often have a noise figure of around
1 dB. However it is interesting to note that even the best professional wide-band VHF
UHF receivers may only have a noise figure of around 8 dB.
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Radio receiver noise floor
- an overview of the noise floor or a receiver, what it is and how the noise
floor affects the performance of a receiver.
Noise is a fact of life. Despite the best efforts of any design engineers, there is always
some background noise present in any radio receiver. The noise emanates from many
sources, and although the design of the receiver is optimised to reduce it some will
always be present.

Accordingly a concept that is very useful in many elements of signal theory and hence in
radio receiver design is that of a noise floor. The noise floor can be defined as the
measure of the signal created from the sum of all the noise sources and unwanted signals
within a system.
In order to reduce the levels of noise and thereby improve the sensitivity of the receiver,
the main element of the receiver that requires its performance to be optimised is the RF
amplifier. The use of a low noise amplifier at the front end of the receiver will ensure that
its performance will be maximised. Wither for use at microwaves or lower frequencies,
this RF amplifier is the chief element in determining the performance of the whole
receiver. The next most important element is the first mixer.
Receiver noise floor
While noise can emanate from many sources, when looking purely at the receiver, the
noise is dependent upon a number of elements. The first is the minimum equivalent input
noise for the receiver. This can be calculated from the following formula:
P = k T B
Where:
P is the power in watts
K is Boltzmann's constant (1.38 x 10^-23 J/K)
B is the bandwidth in Hertz
Using this formula it is possible to determine that the minimum equivalent input noise for
a receiver at room temperature (290K) is -174 dBm / Hz.
It is then possible to calculate the noise floor for the receiver:
Noise floor = -174 + NF + 10 log Bandwidth
Where NF is the noise figure
dBm is the power level expressed in decibels relative to one milliwatt
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Radio receiver strong signal response
- including intermodulation distortion, third order intercept point, cross modulation
and blocking
Receiver sensitivity is important but equally so is the way in which a receiver handles
strong signals. Specifications including intermodulation distortion, third order intercept
point, cross modulation and blocking can be equally vital. In any receiver design a good
balance must be achieved between the sensitivity and the strong signal handling
capability. Under some conditions receivers may need to contend with signals that are
only a few microvolts, but equally they need to handle the conditions when many
millivolts enter the front end.,/p>
RF amplifier
Under normal conditions the RF amplifiers should remain linear with the output
remaining proportional to the input. Unfortunately even the best amplifiers have limits to

their output capability, and beyond this they start to overload. When this happens their
output starts to limit and the output is less than expected. At this point the amplifier is
said to be in compression.
The characteristic curve for an amplifier
Compression in itself is not a problem. The absolute values of a signal are of little value
and in any case the automatic gain control (AGC) used in most receivers means that the
gain is reduced when strong signals are being received. However the side effects of
compression give rise to major problems. Effects like intermodulation distortion, cross
modulation, blocking and others mean that the operation of the receiver can be seriously
impaired. It is these aspects which are of great importance in the receiver design.
To help prevent these problems occurring, receivers have a number of methods of
reducing the signals levels. The most important is the AGC. This is standard on virtually
every receiver and operates on many of the amplifier stages within the set. It prevents the
signals from becoming too large, especially in the later stages of the set. However it
cannot always prevent the front end stages from being overloaded. This is particularly
true when the offending strong signal is slightly off channel. In this case it will enter the
early stages of the set but not pass through the IF filters (assuming the receiver is a
superhet). This will mean that the AGC will not be affected but the signal is still able to
overload some of the early stages.
Some HF communications receivers have an attenuator on the input, although many
receivers used in applications such as cellular telecommunications, PMR and the like will
not have these and the receiver will need to be able to handle the strong signals without
this assistance.
In view of the importance of the various aspects of overloading, a number of
specifications quantify the various problems caused. However to look at these it is
necessary to look at the effects and how they arise.
Distortion
The problems from compression arise as a result of the distortion which occurs to the
signal when the amplifier runs into compression. The actual method which gives rise to
problems may not be obvious at first sight. It can be viewed as the combination of two
effects. However to see how it arises it is necessary to look at some of the basic effects of
compression.
One of the forms of distortion which arises is harmonic distortion where harmonics of the
wanted signal are produced. Depending upon the exact way in which the signal is
compressed the levels of even order harmonics (2f, 4f, 6f, etc) and odd order harmonics
(3f, 5f, 7f, etc) will vary. As a result of the production of these harmonics it is possible
that signals below that being received could be picked up. However the RF selectivity is
likely to remove these signals before they enter the first stages of the receiver.
Another effect which can be noticed is that the amplifier tends to act as a mixer. The nonlinear
transfer curve means that signals will mix together or modulate one another. This
effect is known as intermodulation. It is unlikely that this effect on its own would give
any problems. The mix products from signals close to the wanted one fall well away from

the received signal. Alternatively, to produce a signal within the receiver pass-band,
signals well away from the received one would need to be entering the r.f. amplifier.
These would normally be rejected by the RF selectivity. Take the example of two signals
on 50.00 and 50.01 MHz. These would mix together to give signals at 0.01 MHz and
100.01 MHz. These are not likely to give rise to any problems.
Problems start to arise when the two effects combine with one another. It is quite possible
for a harmonic of one signal to mix with the fundamental or a harmonic of the other. The
third order sum products like 2f1 + f2 are unlikely to cause a problem, but the difference
products like 2f1 - f2 can give significant problems. Take the example of a receiver set to
50 MHz where two strong signals are present, one at 50.00 MHz and the other at 50.01
MHz. The difference signals produced will be at 2 x 50.00 - 50.01 = 49.99 MHz and
another at 2 x 50.01 - 50 = 50.01 MHz. As it can be seen either of these could cause
interference on the band. Other higher order products can also cause problems: 3f1 - 2f2,
4f1 - 3f2, 5f1 - 4f2, and so forth all give products which may could pass through the
receiver if it is tuned to the relevant frequency.
Intermodulation products from two signals
In this way the presence of a strong signal can produce other spurious signals which can
appear in its vicinity. The signals mixing with one another in this way may be of a variety
of different types, e.g. AM, FM, digital modulation, etc, all of which may combine
together to give what is effectively noise. This means that poor third order
intermodulation performance can have the effect of raising the noise floor under real
operating conditions.
Third Order Intercept
It is found that the level of intermodulation products rise very fast. For a 1 dB increase in
wanted signal levels, third order products will rise by 3 dB, and fifth order ones by 5 dB.
This can be plotted to give a graph of the performance of the amplifier. Eventually the
amplifier will run into saturation and the levels of all the signals will be limited. However
if the curve of the wanted signals and the third order products was continued, the two
lines would intersect. This is known as the third order intercept point. Naturally the
higher the level of the intercept point, the better the performance of the amplifier. For a
good receiver and intercept point of 25 dBm (i.e. 25 dB above 1 milliwatt or about 0.5
watt) might be expected.

The third order intercept point of an amplifier
Blocking
When a very strong off channel signal appears at the input to a receiver it is often found
that the sensitivity is reduced. The effect arises because the front end amplifiers run into
compression as a result of the off channel signal. This often arises when a receiver and
transmitter are run from the same site and the transmitter signal is exceedingly strong.
When this occurs it has the effect of suppressing all the other signals trying to pass
through the amplifier, giving the effect of a reduction in gain.
Blocking is generally specified as the level of the unwanted signal at a given offset
(normally 20 kHz) which will give a 3 dB reduction in gain. A good receiver may be able
to withstand signals of about ten milliwatts before this happens.
Cross modulation
Another effect which can be noticed when there are strong signals entering the receiver is
known as cross modulation. When this occurs the modulation from a strong signal can be
transferred onto other signals being picked up. This effect is particularly obvious when
amplitude modulated signals are being received. In this case the modulation of another
signal can be clearly heard.
Cross modulation normally arises out of imperfect mixer performance in the radio,
although it can easily occur in one of the RF amplifiers. As it is a third order effect, a
receiver with a good third order intercept point should also exhibit good cross modulation
performance.
To specify the cross modulation performance the effect of a strong AM carrier on a
smaller wanted signal is noted. Generally the level of a strong carrier with 30%
modulation needed to produce an output 20 dB below that produced by the wanted signal.
The wanted signal level also has to be specified and 1mV or -47dBm (i.e. a signal 47 dB
below 1 mW) is often taken as standard, together with an offset frequency of 20 kHz.
Sensitivity is one of the main specifications of any radio receiver. However the
sensitivity of a set is by no means the whole story. The specification for a set may show it
to have an exceedingly good level of sensitivity, but when it is connected to an antenna
its performance may be very disappointing because it is easily overloaded when strong
signals are present, and this may impair its ability to receive weak signals.
The overall dynamic range of the receiver is very important. It is just as important for a
set to be able to handle strong signals well as it is to be able to pick up weak ones. This
becomes very important when trying to pick up weak signals in the presence of nearby
strong ones. Under these circumstances a set with a poor dynamic range may not be able
to hear the weak stations picked up by a less sensitive set with a better dynamic range.
Problems like blocking, inter-modulation distortion and the like within the receiver may
mask out the weak signals, despite the set having a very good level of sensitivity.
What is dynamic range?
The dynamic range of a receiver is essentially the range of signal levels over which it can
operate. The low end of the range is governed by its sensitivity whilst at the high end it is
governed by its overload or strong signal handling performance. Specifications generally
use figures based on either the inter-modulation performance or the blocking
performance. Unfortunately it is not always possible to compare one set with another
because dynamic range like many other parameters can be quoted in a number of ways.
However to gain an idea of exactly what the dynamic range of a receiver means it is
worth looking at the ways in which the measurements are made to determine the range of
the receiver.
Sensitivity
The first specification to investigate is the sensitivity of a set. The main limiting factor in
any receiver is the noise generated. For most applications either the signal to noise ratio
or the noise figure is used as described in a previous issue of MT. However for dynamic
range specifications a figure called the minimum discernible signal (MDS) is often used.

This is normally taken as a signal equal in strength to the noise level. As the noise level is
dependent upon the bandwidth used, this also has to be mentioned in the specification.
Normally the level of the level of the MDS is given in dBm i.e. dB relative to a milliwatt
and typical values are around -135 dBm in a 3 kHz bandwidth.
Strong signal handling
Although the sensitivity is important the way in which a receiver handles strong signals is
also very important. Here the overload performance governs how well the receiver
performance.
In the ideal world the output of an amplifier would be proportional to the input for all
signal levels. However amplifiers only have a limited output capability and it is found
that beyond a certain level the output falls below the required level because it cannot
handle the large levels required of it. This gives a characteristic like that shown in Fig. 1.
From this it can be seen that amplifiers are linear for the lower part of the characteristic,
but as the output stages are unable to handle the higher power levels the signals starts to
become compressed as seen by the curve in the characteristic.
A typical amplifier characteristic
The fact that the amplifier is non-linear does not create a major problem in itself.
However the side effects do. When a signal is passed through a non-linear element there
are two main effects which are noticed. The first is that harmonics are generated.
Fortunately these are unlikely to cause a major problem. For a harmonic to fall near the
frequency being received, a signal at half the received frequency must enter the amplifier.
The front end tuning should reduce this by a sufficient degree for it not to be a noticeable
problem under most circumstances.
The other problem that can be noticed is that signals mix together to form unwanted
products. These again are unlikely to cause a problem because any signals which could
mix together should be removed sufficiently by the front end tuning. Instead problems
occur when harmonics of in-band signals mix together.
Third order products
Problems occur when harmonics of in-band signals mix together. It is found that a comb
of signals can be produced as shown in Figure 2, and these may just fall on the same
frequency as a weak and intersting station, thereby masking it out so it cannot be heard.
It is simple to calculate the frequencies where the spurious signals will fall. If the input
frequencies are f 1 and f 2 , then the new frequencies produced will be at 2f 1 - f 2 , 3f 1 - 2f 2 ,
4f 1 - 3f 2 and so forth. On the other side of the two main or original signals products are
produced at 2f 2 - f 1 , 3f 2 - 2f 2 , 4f 2 - 3f 1 and so forth as shown in the diagram. These are
known as odd order inter-modulation products. Two times one signal plus one times
another makes a third order product, three times one plus two times another is a fifth
order product and so forth. It can be seen from the diagram that the signals either side of
the main signals are first the third order product, then fifth, seventh and so forth.
To take an example with some real figures. If large signals appear at frequencies of 30.0
MHz and 30.01 MHz, then the inter-modulation products will appear at 30.02, 30.03,
30.4 ...MHz and 29.99, 29.98, 29.97 ..... MHz.

Inter-modulation products
Blocking
Another problem that can occur when a strong signal is present is known as blocking. As
the name implies it is possible for a strong signal to block or at least reduce the sensitivity
of a receiver. The effect can be noticed when listening to a relatively weak station and a
nearby transmitter starts to radiate, and the wanted signal reduces in strength. The effect
is caused when the front-end amplifier starts to run into compression. When this occurs
the strongest signal tends to "capture" the amplifier reducing the strength of the other
signals. The effect is the same as the capture effect associated with FM signals.
The amount of blocking is obviously dependent upon the level of the signal. It also
depends on how far off channel the strong signal is. The further away, the more it will be
reduced by the front end tuning and the less the effect will be. Normally blocking is
quoted as the level of the unwanted signal at a given offset (normally 20 kHz) to give a 3
dB reduction in gain.
Dynamic range definition
When looking at dynamic range specifications, care must be taken when interpreting
them. The MDS at the low signal end should be viewed carefully, but the limiting factors
at the top end show a much greater variation tin the way they are specified. Where
blocking is used a reduction of 3 dB sensitivity is normally specified, but in some cases
may be 1 dB used. Where the inter-modulation products are chosen as the limiting point
the input signal level for them to be the same as the MDS is often taken. However
whatever specification is given, care should be taken to interpret the figures as they may
be subtlety different in the way they are measured from one receiver to the next.
To gain a feel for the figures which may be obtained where inter-modulation is the
limiting factor figures of between 80 and 90 dB range are typical, and where blocking is
the limiting factor figures around 115 dB are generally achieved in a good receiver.
Designing for optimum performance
It is not an easy task to design a highly sensitive receiver that also has a wide dynamic
range. To achieve this performance a number of methods can be used. The front-end
stage is the most critical in terms of noise performance. It should be optimised for noise
performance rather than gain. Input impedance matching is critical for this. It is
interesting to note that the optimum match does not correspond exactly with the best
noise performance. The amplifier should also have a relatively high output capability to
ensure it does not overload. The mixer is also critical to the overload performance. To
ensure the mixer is not overloaded there should not be excessive gain preceding it. A high
level mixer should also be used (i.e. one designed to accept a high-level local oscillator
signal). In this way it can tolerate high input signals without degradation in performance.
Care should be taken in the later stages of the receiver to ensure that they can tolerate the
level of signals likely to be encountered. A good AGC system also helps prevent
overloading and the generation of unwanted spurious signals.
A receiver with a good dynamic range will be able to give a far better account of itself
under exacting conditions than one designed purely for optimum sensitivity.

Frequency modulation is widely used in radio communications and broadcasting,
particularly on frequencies above 30 MHz. It offers many advantages, particularly in
mobile radio applications where its resistance to fading and interference is a great
advantage. It is also widely used for broadcasting on VHF frequencies where it is able to
provide a medium for high quality audio transmissions.
In view of its widespread use, a wide variety of receivers are able to demodulate these
transmissions. Naturally there are specifications and figures that receiver manufacturers
quote for the performance of their sets when receiving FM. These include sch figures as
quieting, capture ratio and the like.
Receiving FM
In order to be able to receive FM a receiver must be sensitive to the frequency variations
of the incoming signals which may be wide or narrow band. However the set is made
insensitive to the amplitude variations. This is achieved by having a high gain IF
amplifier. Here the signals are amplified to such a degree that the amplifier runs into
limiting. In this way any amplitude variations are removed and this improves the signal to
noise ratio after the point when the signal limits in the IF stages. However the high levels
of gain associated with the limiting process mean that when no signal is present, very
high levels of noise appear at the output of the FM demodulator.,/p>
Squelch
To overcome the problem of the high noise levels when no signal is present a circuit
known as "squelch" is normally used. This detects when no signal is present and cuts the
audio, thereby removing the noise under these conditions. The level for this is normally
present in domestic radios, but there is often a level adjustment for PMR or handheld
transceivers, or for scanners and professional receivers.
Quieting specification
One of the advantages of FM is its resilience to noise. This is one of the main reasons
why it is used for high quality audio broadcasts. However when no signal is present, a
high noise level is present at the output of the receiver. If a low level FM signal is
introduced and its level slowly increased it will be found that the noise level reduces.
From this the quieting level can be deduced. It is the reduction in noise level expressed in
decibels when a signal of a given strength is introduced to the input of the set. Typically a
broadcast tuner should give a quieting level of 30 dB for an input level of around a
microvolt.
Capture effect
Another effect that is often associated with FM is called the capture effect. This can be
demonstrated when two signals are present on the same frequency. When this occurs it is
found that only the stronger signal will heard at the output This can be compared to AM
where a mixture of the two signals is heard, along with a heterodyne if there is a
frequency difference.
A capture ratio is often defined in receiver specifications. It is the ratio between the
wanted and unwanted signal to give a certain reduction in level of the unwanted signal at
the output. Normally a reduction of the unwanted signal of 30 dB is used. To give an
example of this the capture ratio may be 2 dB for a typical tuner to give a reduction of 30
dB in the unwanted signal. In other words if the wanted signal is only 2 dB stronger than
the unwanted one, the audio level of the unwanted one will be suppressed by 30 dB.
The phase locked loop or PLL is a particularly flexible circuit building block. The phase
locked loop, PLL can be used for a variety of radio frequency applications, and
accordingly the PLL is found in many radio receivers as well as other pieces of
equipment.
The phase locked loop, PLL, was not used in early radio equipment because of the
number of different stages required. However with the advent of radio frequency
integrated circuits, the idea of phase locked loops, PLLs, became viable. Initially
relatively low frequency PLLs became available, but as RF IC technology improved, so

the frequency at which PLLs would operate rose, and high frequency versions became
available.
Phase locked loops are used ain a large variety of applications within radio frequency
technology. PLLs can be used as FM demodulators and they also form the basis of
indirect frequency synthesizers. In addition to this they can be used for a number of
applications including the regeneration of chopped signals such as the colour burst signal
on an analogue colour television signal, for types of variable frequency filter and a host
of other specialist applications
Concepts - phase
The operation of a phase locked loop, PLL, is based around the idea of comparing the
phase of two signals. This information about the error in phase or the phase difference
between the two signals is then used to control the frequency of the loop.
To understand more about the concept of phase and phase difference, first visualise a
radio frequency signal in the form of a familiar x-y plot of a sine wave. As time
progresses the amplitude oscillates above and below the line, repeating itself after each
cycle. The linear plot can also be represented in the form of a circle. The beginning of the
cycle can be represented as a particular point on the circle and as a time progresses the
point on the waveform moves around the circle. Thus a complete cycle is equivalent to
360 degrees. The instantaneous position on the circle represents the phase at that given
moment relative to the beginning of the cycle.
To look at the concept of phase difference, take the example of two signals. Although the
two signals have the same frequency, the peaks and troughs do not occur in the same
place. There is said to be a phase difference between the two signals. This phase
difference is measured as the angle between them. It can be seen that it is the angle
between the same point on the two waveforms. In this case a zero crossing point has been
taken, but any point will suffice provided that it is the same on both.
When there two signals have different frequencies it is found that the phase difference
between the two signals is always varying. The reason for this is that the time for each
cycle is different and accordingly they are moving around the circle at different rates. It
can be inferred from this that the definition of two signals having exactly the same
frequency is that the phase difference between them is constant. There may be a phase
difference between the two signals. This only means that they do not reach the same point
on the waveform at the same time. If the phase difference is fixed it means that one is
lagging behind or leading the other signal by the same amount, i.e. they are on the same
frequency.
PLL basics
A phase locked loop, PLL, is basically of form of servo loop. Although a PLL performs
its actions on a radio frequency signal, all the basic criteria for loop stability and other
parameters are the same.
A basic phase locked loop, PLL, consists of three basic elements:
• Phase comparator: As the name implies, this circuit block within the PLL
compares the phase of two signals and generates a voltage according to the phase
difference between the two signals.
• Loop filter: This filter is used to filter the output from the phase comparator in
the PLL. It is used to remove any components of the signals of which the phase is
being compared from the VCO line. It also governs many of the characteristics of
the loop and its stability.
• Voltage controlled oscillator (VCO): The voltage controlled oscillator is the
circuit block that generates the output radio frequency signal. Its frequency can be
controlled and swung over the operational frequency band for the loop.
PLL operation
The concept of the operation of the PLL is relatively simple, although the mathematical
analysis can become more complicated

The Voltage Controlled Oscillator, VCO, within the PLL produces a signal which enters
the phase detector. Here the phase of the signals from the VCO and the incoming
reference signal are compared and a resulting difference or error voltage is produced.
This corresponds to the phase difference between the two signals.
Block diagram of a basic phase locked loop (PLL)
The error signal from the phase detector in the PLL passes through a low pass filter
which governs many of the properties of the loop and removes any high frequency
elements on the signal. Once through the filter the error signal is applied to the control
terminal of the VCO as its tuning voltage. The sense of any change in this voltage is such
that it tries to reduce the phase difference and hence the frequency between the two
signals. Initially the loop will be out of lock, and the error voltage will pull the frequency
of the VCO towards that of the reference, until it cannot reduce the error any further and
the loop is locked.
When the PLL is in lock a steady state error voltage is produced. By using an amplifier
between the phase detector and the VCO, the actual error between the signals can be
reduced to very small levels. However some voltage must always be present at the
control terminal of the VCO as this is what puts onto the correct frequency.
The fact that a steady error voltage is present means that the phase difference between the
reference signal and the VCO is not changing. As the phase between these two signals is
not changing means that the two signals are on exactly the same frequency.
Summary
The phase locked loop is one of the most versatile building blocks in radio frequency
electronics today. Whilst it was not widely used for many years, the advent of the IC
meant that phase locked loop and synthesizer chips became widely available. This made
them cheap to use and their advantages could be exploited to the full. Nowadays most hifi
tuners and car radios use them and a large proportion of the portable radios on the
market as well. With their interface to microprocessors so easy their use is assured for
many years to come.
The phase detector is the core element of a phase locked loop, PLL. Its action enables the
phase differences in the loop to be detected and the resultant error voltage to be produced.
There is a variety of different circuits that can be used as phase detectors, some that use
what may be considered as analogue techniques, while others use digital circuitry.
However the most important difference is whether the phase detector is sensitive to just
phase or whether it is sensitive to frequency and to phase. Thus phase detectors may be
split into two categories:
• Phase only sensitive detectors
• Phase - frequency detectors
Phase only sensitive detectors
Phase detectors that are only sensitive to phase are the most straightforward form of
detector. They simply produce an output that is proportional to the phase difference
between the two signals. When the phase difference between the two incoming signals is
steady, they produce a constant voltage. When there is a frequency difference between
the two signals, they produce a varying voltage. In fact the simplest form of phase only

sensitive detector is a mixer. From this it can be seen that the output signal will be have
sum and difference signals.
The difference frequency product is the one used to give the phase difference. It is quite
possible that the difference frequency signal will fall outside the pass-band of the loop
filter. If this occurs then no error voltage will be fed back to the Voltage Controlled
Oscillator (VCO) to bring it into lock. This means that there is a limited range over which
the loop can be brought into lock, and this is called the capture range. Once in lock the
loop can generally be pulled over a much wider frequency band.
To overcome this problem the oscillator must be steered close to the reference oscillator
frequency. This can be achieved in a number of ways. One is to reduce the tuning range
of the oscillator so that the difference product will always fall within the pass-band of the
loop filter. In other instances another tune voltage can be combined with the feedback
from the loop to ensure that the oscillator is in the correct region. This is approach is
often adopted in microprocessor systems where the correct voltage can be calculated for
any given circumstance.
Phase - frequency sensitive detectors
Another form of detector is said to be phase-frequency sensitive. These circuits have the
advantage that whilst the phase difference is between +/- 180 a voltage proportional to
the phase difference is given. Beyond this the circuit limits at one of the extremes. In this
way no AC component is produced when the loop is out of lock and the output from the
phase detector can pass through the filter to bring the phase locked loop, PLL, into lock.
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Voltage controlled oscillator, VCO, for PLLs
- an overview of the various types of voltage controlled oscillator, VCO,
used in phase locked loops, PLLs and frequency synthesizers
Within a phase locked loop, PLL, or frequency synthesizer, the performance of the
voltage controlled oscillator, VCO is of paramount importance. In order that the PLL or
synthesizer can meet its full specification a well designed oscillator is essential.
Designing a really high performance voltage controlled oscillator, VCO, is not always
easy as there are a number of requirements that need to be met. However by careful
design, and some experimentation a good VCO design can be found.
VCO requirements
Just like any other circuit, with a VCO there are a number of design requirements that
need to be known from the beginning of the design process. These basic requirements for
the VCO will govern many of the decisions concerning the circuit topology and other
fundamental aspects of the circuit. Some of the basic requirements are:
• Tuning range
• Tuning gain - tuning shift for a given tuning voltage change
• Phase noise (low phase noise)
These are some of the main requirements that need to be known from the outset of the
design of the VCO. The overall tuning range and the gain are basic requirements that are
part of the basic design of any PLL into which the VCO may be incorporated. So too is

the phase noise characteristic. As phase noise is a basic parameter of any PLL or
frequency synthesizer, so too is the characteristic of the VCO, and low phase noise VCOs
are often required. For example the VCO performance may govern the overall design of
the frequency synthesizer or PLL, if a given phase noise performance is to be met.
VCO circuits
Like any oscillator, a VCO may be considered as an amplifier and a feedback loop. The
gain of the amplifier may be denoted as A and the feedback as B.
For the circuit to oscillate the total phase shift around the loop must be 360 degrees and
the gain must be unity. In this way signals are fed back round the loop so that they are
additive and as a result, any small disturbance in the loop is fed back and builds up. In
view of the fact that the feedback network is frequency dependent, the build up of signal
will occur on one frequency, the resonant frequency of the feedback network, and a
single frequency signal is produced.
Many oscillators and hence VCOs use a common emitter circuit. This in itself produces a
phase shift of 180 degrees, leaving the feedback network to provide a further 180
degrees.
Other oscillator or VCO circuits may use a common base circuit where there is no phase
shift between the emitter and collector signals (assuming a bipolar transistor is used) and
the phase shift network must provide either 0 degrees or 360 degrees.
Colpitts and Clapp VCO circuits
Two commonly used examples of VCO circuits are the Colpitts and Clapp oscillators. Of
the two, the Colpitts circuit is the most widely used, but these circuits are both very
similar in their configuration.
These circuits operate as oscillators because it is found that a bipolar transistor with
capacitors placed between the base and emitter (C1) and the emitter and ground (C2)
fulfils the criteria required for providing sufficient feedback in the correct phase to
produce an oscillator. For oscillation to take place the ratio C1: C2 must be greater than
one.
The resonant circuit is made by including a inductive element between the base and
ground. In the Colpitts circuit this consists of just an inductor, whereas in the Clapp
circuit an indictor and capacitor in series are used.
The conditions for resonance is that:
f^2 = 1 / (4 pi^2 L C )
The capacitance for the overall resonant circuit is formed by the series combination of the
two capacitors C1 and C2 in series. In the case of the Clapp oscillator, the capacitor in
series with the inductor is also included in series with C1 and C2.
Thus the series capacitance is:
Ctot
= 1 / C1 + 1 / C2
In order to make the oscillator tune it is necessary to vary the resonant point of the circuit.
This is best achieved by adding a capacitor across the indictor in the case of the Colpitts
oscillator. Alternatively for the Clapp oscillator, it can be the capacitor in series with the
inductor.
For high frequency applications a circuit where the inductive reactance is placed between
the base and ground is often preferred as it is less prone to spurious oscillations and other
anomalies.
Choice of VCO active device
It is possible to use both bipolar devices and FETs within a VCO, using the same basic

circuit topologies. The bipolar transistor has a low input impedance and is current driven,
while the FET has a high input impedance and is voltage driven. The high input
impedance of the FET is able to better maintain the Q of the tuned circuit and this should
give a better level of performance in terms of the phase noise performance where the
maintenance of the Q of the tuned circuit is a key factor in the reduction of phase noise.
Another major factor is the flicker noise generated by the devices. Oscillators are highly
non-linear circuits and as a result the flicker noise is modulated onto VCO as sidebands
and this manifests itself as phase noise. In general bipolar transistors offer a lower level
of flicker noise and as a result VCOs based around them offer a superior phase noise
performance.
VCO tuning
To make a VCO, the oscillator needs to be tuned by a voltage. This can be achieved by
making the variable capacitor from varactor diodes. The tune voltage for the VCO can
then be applied to the varactors.
When varactor diodes are used, care must be taken in the design of the circuit to ensure
that the drive level in the tuned circuit is not too high. If this is the case, then the varactor
diodes may be driven into forward conduction, reducing the Q and increasing the level of
spurious signals.
There are two main types of varactor diode that may be used within a VCO: abrupt and
hyper-abrupt diodes. The names refer to the junction within the diode. The abrupt ones do
not have a sharp a transition between the two semiconductor types in the diode, and this
affect the performance offered.
Hyper-abrupt diodes have a relatively linear voltage : capacitance curve and as a result
they offer a very linear tuning characteristic that may be required in some applications.
They are also able to tune over a wide range, and may typically tune over an octave range
with less than a 20 volt change in tuning voltage. However they do not offer a
particularly high level of Q. As this will subtract from the overall Q of the tuned circuit
this will mean that the phase noise performance is not optimum.
Abrupt diodes, while not offering such a high tuning range or linear transfer characteristic
are able to offer a higher Q. This results in a better phase noise (i.e. low phase noise)
performance for the VCO. The other point to note is that they may need a high tuning
voltage to provide the required tuning range, as some diodes may require a tuning voltage
for the VCO to vary up to 50 volts or slightly more.
Summary
The design of a VCO can be interesting and challenging. Whether the aim is to design a
low noise VCO, a low current VCO, a PLL VCO, or one that will cover a wide tuning
range there are many aspects that need to be addressed. Often when a successful design
has been obtained, it will slightly modified to enable it to cover a wide range of similar
applications.
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Oscillator design for low phase noise
- an overview of the design of radio frequency, RF, oscillators for low
phase noise levels.
One of the key requirements for any oscillator used in a radio receiver, radio transmitter,
or many other applications is for the oscillator to perform with low levels of phase noise.
Whether the oscillator is used in a frequency synthesizer, or in any other application, the
basic design principles for achieving low phase noise are the same.

Poor levels of oscillator can manifest themselves in slightly different ways. For an
analogue radio receiver a poor performance oscillator may result in poor reciprocal
mixing performance. It may also raise the noise floor of the receiver. In a radio system
relying on phase modulation, phase noise will degrade the bit error rate performance.
Additionally transmitters exhibiting a poor phase noise performance will tend to transmit
wide band noise, causing interference to users on other frequencies.
Key points for oscillator design
There are some areas points to address when designing an oscillator to ensure that it has a
good phase noise performance. By addressing these and other relevant points, the
performance of the oscillator meets its requirements.
• High Q resonant circuit
• Choice of oscillator device
• Correct feedback level
• Sufficient oscillator power output
• Power line rejection
Oscillator design methodology
In order that the oscillator is able to provide the optimum phase noise performance it is
necessary to implement these elements into the design of the circuit from the outset.
Some aspects may affect the basic design criteria, and therefore need to be included from
the concept stage of the circuit:
Q of the resonant circuit: One of the major factors in determining the phase noise
performance of an oscillator is the Q of the resonant circuit. Broadly, the higher the Q of
the oscillator tuned circuit, the better the phase noise performance. This inductors should
be chosen to provide the highest Q, as should the capacitors. This is particularly true of
voltage controlled oscillators, VCOs where the varactor diodes normally employed have
a lower Q than other capacitors.
Typically high Q tuned circuits do not have the tuning range of lower Q circuits. This
means that when wide tuning ranges are required, it becomes more difficult to obtain a
high level of Q and hence the optimum phase noise.
As an illustration of the effect of having a high Q resonant circuit in an oscillator, crystal
oscillators exhibit very low levels of phase noise as a result of the fact that the crystals
used in them possess very high levels of Q.
Choice of oscillator active device: It is possible to use both bipolar devices and FETs
within an RF oscillator, using the same basic circuit topologies. The bipolar transistor has
a low input impedance and is current driven, while the FET has a high input impedance
and is voltage driven. The high input impedance of the FET is able to better maintain the
Q of the tuned circuit and this should give a better level of performance in terms of the
phase noise performance where the maintenance of the Q of the tuned circuit is a key
factor in the reduction of phase noise.

Another major factor is the flicker noise generated by the devices. Oscillators are highly
non-linear circuits and as a result the flicker noise is modulated onto the oscillation as
sidebands. This manifests itself as phase noise. In general bipolar transistors offer a lower
level of flicker noise and as a result oscillators based around them offer a superior phase
noise performance.
Oscillator feedback level: A critical feature in any oscillator design is to ensure that the
correct level of feedback is maintained. There should be sufficient to ensure that
oscillation is maintained over the frequency range, over the envisaged temperature range
and to accommodate the gain and parameter variations between the devices used.
However if the level of feedback is too high, then the level of noise will also be
increased. Thus the circuit should be designed to provide sufficient feedback for reliable
operation and little more.
Sufficient oscillator power output: It is found that the noise floor of an oscillator is
reasonably constant in absolute terms despite the level of the output signal. In some
designs there can be improvements in the overall signal to noise floor level to be made by
using a high level signal and applying this directly to the mixer or other circuit where it
may be required. Accordingly some low noise circuits may use surprisingly high
oscillator power levels.
Power line rejection: It is necessary to ensure that any supply line or other extraneous
noise is not presented to the oscillator. Supply line ripple, or other unwanted pickup can
seriously degrade the performance of the oscillator. To overcome this, good supply
smoothing and regulation is absolutely necessary. Additionally it may be advisable to
place the oscillator within a screened environment so that it does not pick up any stray
noise. It is worth remembering that the oscillator acts as a high gain amplifier, especially
close to the resonant frequency. Any noise picked up can be amplified and will manifest
itself as phase noise.
Summary
There are many elements to ensuring that an oscillator circuit design meets its
requirements for low phase noise. The points provider here give a start to some of the
basic decisions that are needed. Once initially realised, some refinement is likely to be
needed to ensure the optimum performance is obtained.
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PLL loop filter
- an overview of the loop filter used in a phase locked loop, PLL. This gives
an overview of the requirements, and design.
The design of the PLL, loop filter is crucial to the operation of the whole phase locked
loop. The choice of the circuit values here is usually a very carefully balanced
compromise between a number of conflicting requirements.
The PLL filter is needed to remove any unwanted high frequency components which
might pass out of the phase detector and appear in the VCO tune line. They would then
appear on the output of the Voltage Controlled Oscillator, VCO, as spurious signals. To
show how this happens take the case when a mixer is used as a phase detector. When the
loop is in lock the mixer will produce two signals: the sum and difference frequencies. As
the two signals entering the phase detector have the same frequency the difference
frequency is zero and a DC voltage is produced proportional to the phase difference as
expected. The sum frequency is also produced and this will fall at a point equal to twice
the frequency of the reference. If this signal is not attenuated it will reach the control
voltage input to the VCO and give rise to spurious signals.
When other types of phase detector are used similar spurious signals can be produced and
the filter is needed to remove them.

The filter also affects the ability of the loop to change frequencies quickly. If the filter
has a very low cut-off frequency then the changes in tune voltage will only take place
slowly, and the VCO will not be able to change its frequency as fast. This is because a
filter with a low cut-off frequency will only let low frequencies through and these
correspond to slow changes in voltage level.
Conversely a filter with a higher cut-off frequency will enable the changes to happen
faster. However when using filters with high cut-off frequencies, care must be taken to
ensure that unwanted frequencies are not passed along the tune line with the result that
spurious signals are generated.
The loop filter also governs the stability of the loop. If the filter is not designed correctly
then oscillations can build up around the loop, and large signals will appear on the tune
line. This will result in the VCO being forced to sweep over wide bands of frequencies.
The proper design of the filter will ensure that this cannot happen under any
circumstances.
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PLL Frequency synthesizer tutorial
- an introduction to the indirect (phase locked loop - pll) synthesizer
Today most receivers use a phase locked loop or PLL frequency synthesizer. Many of
them advertise this fact by displaying words like "PLL", "Synthesized", or "Quartz" on
their front panels or in the advertising literature. Whatever one thinks of the sales
language, PLL frequency synthesizers offer tremendous advantages to the operation of a
receiver. Not only do frequency synthesizers enable receivers to have the same stability
as the quartz reference, but they also enable many other facilities to be introduced
because they can easily be controlled by a microprocessor. This enables facilities such as
multiple memories, keypad frequency entry, scanning and much more to be incorporated
into the set.
Phase locked loop, PLL, frequency synthesizers are widely used, but their operation is
not always well understood. One of the reasons for this is that their design can involve
some complicated math, but despite this the basic concepts are relatively easy to grasp.
PLL Basics
A frequency synthesizer is based around a phase locked loop or PLL. This circuit uses the
idea of phase comparison as the basis of its operation. From the block diagram of a basic
loop shown in Fig. 1 it can be seen that there are three basic circuit blocks, a phase
comparator, voltage controlled oscillator, and loop filter. A reference oscillator is
sometimes included in the block diagram, although this is not strictly part of the loop
itself even though a reference signal is required for its operation.
Block diagram of a basic phase locked loop (PLL)
The phase locked loop, PLL, operates by comparing the phase of two signals. The signals
from the voltage controlled oscillator and reference enter the phase comparator Here a
third signal equal to the phase difference between the two input signals is produced.

The phase difference signal is then passed through the loop filter. This performs a
number of functions including the removal of any unwanted products that are present on
this signal. Once this has been accomplished it is applied to the control terminal of the
voltage controlled oscillator. This tune voltage or error voltage is such that it tries to
reduce the error between the two signals entering the phase comparator. This means that
the voltage controlled oscillator will be pulled towards the frequency of the reference,
and when in lock there is a steady state error voltage. This is proportional to the phase
error between the two signals, and it is constant. Only when the phase between two
signals is changing is there a frequency difference. As the phase difference remains
constant when the loop is in lock this means that the frequency of the voltage controlled
oscillator is exactly the same as the reference.
Synthesisizers
A phase locked loop, PLL, needs some additional circuitry if it is to be converted into a
frequency synthesizer. This is done by adding a frequency divider between the voltage
controlled oscillator and the phase comparator as shown in Fig. 2.
A programmable divider added into a phase locked loop, PLL, enables the
frequency to be changed.
Programmable dividers or counters are used in many areas of electronics, including many
radio frequency applications. They take in a pulse train like that shown in Fig. 3, and give
out a slower train. In a divide by two circuit only one pulse is given out for every two that
are fed in and so forth. Some are fixed, having only one division ratio. Others are
programmable and digital or logic information can be fed into them to set the division
ratio.
Operation of a programmable divider
When the divider is added into the circuit the phase locked loop, PLL, still tries to reduce
the phase difference between the two signals entering the phase comparator. Again
when the circuit is in lock both signals entering the comparator are exactly the same in
frequency. For this to be true the voltage controlled oscillator must be running at a
frequency equal to the phase comparison frequency times the division ratio.
It can be seen that if the division ratio is altered by one, then the voltage controlled
oscillator will have to change to the next multiple of the reference frequency. This means
that the step frequency of the synthesizer is equal to the frequency entering the
comparator.
Most synthesizers need to be able to step in much smaller increments if they are to be of
any use. This means that the comparison frequency must be reduced. This is usually

accomplished by running the reference oscillator at a frequency of a megahertz or so, and
then dividing this signal down to the required frequency using a fixed divider. In this way
a low comparison frequency can be achieved.
Comparison frequency reduced by adding a fixed divider after the reference
oscillator
Analogue Techniques
Placing a digital divider is not the only method of making a synthesizer using a phase
locked loop, PLL. It is also possible to use a mixer in the loop as shown in Fig. 5. Using
this technique places an offset into the frequency generated by the loop.
A phase locked loop, PLL, with mixer
The way in which the phase locked loop, PLL, operates with the mixer incorporated can
be analyzed in the same manner that was used for the loop with a divider. When the loop
is in lock the signals entering the phase detector are at exactly the same frequencies. The
mixer adds an offset equal to the frequency of the signal entering the other port of the
mixer. To illustrate the way this operates figures have been included. If the reference
oscillator is operating at a frequency of 10 MHz and the external signal is at 15 MHz then
the VCO must operate at either 5 MHz or 25 MHz.. Normally the loop is set up so that
mixer changes the frequency down and if this is the case then the oscillator will be
operating at 25 MHz.
It can be seen that there may be problems with the possibility of two mix products being
able to give the correct phase comparison frequency. It happens that as a result of the
phasing in the loop, only one will enable it to lock. However to prevent the loop getting
into an unwanted state the range of the VCO is limited. For phase locked loops, PLLs,
that need to operate over a wide range a steering voltage is added to the main tune
voltage so that the frequency of the loop is steered into the correct region for required
conditions. It is relatively easy to generate a steering voltage by using digital information
from a microprocessor and converting this into an analogue voltage using a digital to

analogue converter (DAC). The fine tune voltage required to pull the loop into lock is
provided by the loop in the normal way.
Multi-loop synthesizers
Many high performance synthesizers use several loops that incorporate both mixers and
digital dividers. By using these techniques it is possible to produce high performance
wide range signal sources with very small step sizes. If only a single loop is used then
there may be short falls in the level of performance.
There is a large variety of ways in which multi-loop synthesizers can be made, dependent
upon the requirements of the individual system. However as an illustration a two loop
system is shown in Fig. 6. This uses one loop to give the smaller steps and the second
provides larger steps. This principle can be expanded to give wider ranges and smaller
steps.
An example of a synthesizer using two loops
The first phase locked loop, PLL, has a digital divider and operates over the range 19 to
28 MHz. Having a reference frequency of 1 MHz it provides steps of 1 MHz. The signal
from this loop is fed into the mixer of the second one. The second loop has division ratios
of 10 to 19, but as the reference frequency has been divided by 10 to 100 kHz to give
smaller steps.

The operation of the whole loop can be examined by looking at extremes of the
frequency range. With the first loop set to its lowest value the divider is set to 19 and the
output from the loop is at 19 MHz. This feeds into the second loop. Again this is set to
the minimum value and the frequency after the mixer must be at 1.0 MHz. With the input
from the first loop at 19 MHz this means that the VCO must operate at 20 MHz if the
loop is to remain in lock.
At the other end of the range the divider of the first loop is set to 28, giving a frequency
of 28 MHz. The second phase locked loop, PLL, has the divider set to 19, giving a
frequency of 1.9 MHz between the mixer and divider. In turn this means that the
frequency of the VCO must operate at 29.9 MHz. As the phase locked loops, PLLs, can
be stepped independently it means that the whole synthesizer can move in steps of 100
kHz between the two extremes of frequency. As mentioned before this principle can be
extended to give greater ranges and smaller steps, providing for the needs of modern
receivers.
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Phase noise and frequency synthesizers
One of the main problems with frequency synthesisers and frequency synthesis using
phase locked loops is the fact that some designs generate high levels of phase noise.
However it is possible to design some very good low phase noise synthesizers. The
problem is often that receivers and transceivers are designed for low production costs,
and this naturally means that some short cuts are needed.
What is phase noise?
Phase noise is present on all signals to some degree and it is caused by small phase (and
hence frequency) perturbations or jitter on the signal. It manifests itself as noise
spreading out either side from the main carrier
Phase noise on a signal
Some signal sources are better than others. Crystal oscillators are very good and have
very low levels of phase noise. Free running variable frequency oscillators normally
perform well. Unfortunately synthesizers, and especially those based around phase locked
loops, do not always fare so well unless they are well designed. If significant levels of
phase noise are present on a synthesizer used as a local oscillator in a receiver, it can
adversely affect the performance of the radio in terms of reciprocal mixing.
Some oscillators have phase noise levels that are quoted in their specifications. Any high
quality signal generator will have the level of phase noise specified, as do many high
performance crystal oscillators used as standards. Their performance is generally
specified in dBc/Hz and at a given offset from the carrier. The term dBc simply refers to
the level of noise relative to the carrier, i.e. -10 dBc means that the level is 10 lower than
the carrier.

The bandwidth in which the noise is measured also has to be specified. The reason for
this is that noise spreads over the frequency spectrum. Obviously the wider bandwidth
that is used, the greater the level of noise that will pass through the filter and be
measured. To prove this, just try selecting a different bandwidth on a receiver and check
what happens to the noise level. It will rise for a wider bandwidth and fall when a narrow
bandwidth is used. Technically the most convenient bandwidth to use a 1 Hz bandwidth
and so this is used. When measuring this a wider bandwidth is usually used because it is
difficult to obtain 1 Hz bandwidth filters and a correction is made mathematically.
Finally the level of noise varies as different offsets from the carrier are taken.
Accordingly this must be included in a specification. A very good oscillator might have a
specification of -100 dBc/Hz at 10 kHz offset.
It has already been mentioned that the level of phase noise changes as the offset from the
carrier changes and for "simple" signal sources like crystal oscillators or variable
frequency oscillators the phase noise reduces as the frequency from the main carrier is
increased. For frequency synthesizers the picture is a little more complicated as we shall
see.
Phase noise in synthesizers
Each of the components in a frequency synthesizer produces noise that will contribute to
the overall noise that appears at the output. The actual way in which the noise from any
one element in the loop contributes to the output will depend upon where it is produced.
Noise generated by the VCO will affect the output in a different way to that generated in
the phase detector for example.
To see how this happens take the example of noise generated by the voltage controlled
oscillator. This will pass through the divider chain and appear at the output of the phase
detector. It will then have to pass through the loop filter. This will only allow through
those components of the noise that are below the loop cut-off frequency. These will
appear on the error voltage and have the effect of cancelling out the noise on the voltage
controlled oscillator. As this effect will only take place within the loop bandwidth, it will
reduce the level of noise within the loop bandwidth and have no effect on noise outside
the loop bandwidth.
Noise generated by the phase detector is affected in a different way. Again only the
components of the noise below the loop bandwidth will pass through the low pass filter.
This means that there will be no components outside the loop bandwidth appearing on the
tune voltage at the control terminal of the voltage controlled oscillator, and there will be
no effect on the oscillator. Those components inside the loop bandwidth will appear at
the oscillator control terminal. These will affect the oscillator and appear as phase noise
on the output of the voltage controlled oscillator.
Matters are made worse by the fact that the division ratio has the effect of multiplying the
noise level. This arises because the synthesizer effectively has the effect of multiplying
the frequency of the reference. Consequently the noise level is also multiplied by a factor
of 20 log N, where N is the division ratio.
Noise generated by the reference undergoes exactly the same treatments as that generated
by the phase detector. It too is multiplied by the division ratio of the loop in the same way
that the phase detector noise is. This means that even though the reference oscillator may
have a very good phase noise performance this can be degraded significantly, especially
if division ratios are high.
Dividers normally do not produce a significant noise contribution. Any noise they
produce may be combined with that of the phase detector.
The combined noise of the loop at the output generally looks like that shown in Figure 2.
Here it can be seen that the noise within the loop bandwidth arises from the phase
detector and the reference. Outside the loop bandwidth it arises primarily from the
voltage controlled oscillator. From this it can be seen that optimisation of the noise
profile is heavily dependent upon the choice of the loop bandwidth. It is also necessary to

keep the division ratio in any loop down to reasonable levels. For example a 150 MHz
synthesizer with a 12.5 kHz step size will require a division ratio of 12000. In turn this
will degrade the phase detector and reference phase noise figures by 81 dB inside the
loop bandwidth - a significant degradation by anyone's standards! Provided that division
ratios are not too high then a wide loop bandwidth can help keep the voltage controlled
oscillator noise levels down as well.
Noise profile of a typical synthesizer
Effects of phase noise
Phase noise can have a number of effects. For SSB transmitters like those used for HF
communications for ship to shore, amateur radio and other applications the main effect is
that splatter appears either side of the main signal. This results from the phase noise
either side of the signal will rising and falling in line with the amplitude variations of the
main signal. For digital transmissions using frequency or phase modulation, the noise can
introduce errors causing the bit error rate (BER) to rise.
For receivers the main problem is an effect known as reciprocal mixing. To look at how
this occurs take the case of a superhet receiver tuned to a strong signal. The signal will
pass through the radio frequency stages, and then in the mixer it will be mixed with the
local oscillator to produce a new signal at the right frequency to pass through the IF
filters. When the local oscillator is tuned away by ten kilohertz, for example the signal
will no longer be able to pass through the IF filters. However it will still be possible for
the phase noise on the local oscillator to mix with the strong incoming signal to produce a
signal that will fall inside the receiver pass-band as shown in Figure 3. This could be
sufficiently strong to mask out a weak station.
The way in which phase noise on a signal results in reciprocal mixing
Specifications
A number of different methods are used to define the level of reciprocal mixing.
Generally they involve the response of the receiver to a large off channel signal. To
perform a reciprocal mixing measurement is rarely easy. The signal generator must
always be much better than the receiver, otherwise the performance of the signal
generator will be measured! To overcome this many people use an old valve generator
because their performance is often very good in this respect.

A measurement can be made by noting the level of audio with a BFO on from a small
signal. The signal is then tuned off channel by a given amount, normally about 20 kHz
and then increased until the audio level rises to the same level as a result of the phase
noise from the receiver. As the noise level is dependent upon the bandwidth of the
receiver this has to be specified as well. Generally a bandwidth useable for SSB is used
i.e. 2.7 kHz.
For example a good HF communications receiver might have a figure of 95 dB at a 20
kHz offset using a 2.7. kHz bandwidth. This figure will improve as the frequency offset
from the main channel is increased. At 100 kHz one might expect to see a figure in
excess of 105 dB or possibly more.
Another way of measuring the phase noise response is to inject a large signal into the
receiver and monitor the level needed to give a 3 dB increase in background noise level.
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Direct digital synthesis (DDS)
Direct digital synthesis (DDS) is a powerful technique used in the generation of radio
frequency signals for use in a variety of applications from radio receivers to signals
generators and many more. The technique has become far more widespread in recent
years with the advances being made in integrated circuit technology that allow much
faster speeds to be handled which in turn enable higher frequency DDS chips to be made.
Although often used on its own, Direct Digital Synthesis is often used in conjunction
with indirect or phase locked loop synthesizer loops. By combining both technologies it
is possible to take advantage of the best aspects of each. In view of the fact that integrated
circuits are now widely available, this makes them easy to use.
How it works
As the name suggests this form of synthesis generates the waveform directly using digital
techniques. This is different to the way in which the more familiar indirect synthesizers
that use a phase locked loop as the basis of their operation.
A direct digital synthesizer operates by storing the points of a waveform in digital format,
and then recalling them to generate the waveform. The rate at which the synthesizer
completes one waveform then governs the frequency. The overall block diagram is
shown below, but before looking at the details operation of the synthesizer it is necessary
to look at the basic concept behind the system.
The operation can be envisaged more easily by looking at the way that phase progresses
over the course of one cycle of the waveform. This can be envisaged as the phase
progressing around a circle. As the phase advances around the circle, this corresponds to
advances in the waveform.
Block Diagram of a Basic Direct Digital Synthesizer (DDS).
The synthesizer operates by storing various points in the waveform in digital form and
then recalling them to generate the waveform. Its operation can be explained in more

detail by considering the phase advances around a circle as shown in Figure 2. As the
phase advances around the circle this corresponds to advances in the waveform, i.e. the
greater the number corresponding to the phase, the greater the point is along the
waveform. By successively advancing the number corresponding to the phase it is
possible to move further along the waveform cycle.
The digital number representing the phase is held in the phase accumulator. The number
held here corresponds to the phase and is increased at regular intervals. In this way it can
be sent hat the phase accumulator is basically a form of counter. When it is clocked it
adds a preset number to the one already held. When it fills up, it resets and starts counting
from zero again. In other words this corresponds to reaching one complete circle on the
phase diagram and restarting again.
Operation of the phase accumulator in a direct digital synthesizer.
Once the phase has been determined it is necessary to convert this into a digital
representation of the waveform. This is accomplished using a waveform map. This is a
memory which stores a number corresponding to the voltage required for each value of
phase on the waveform. In the case of a synthesizer of this nature it is a sine look up table
as a sine wave is required. In most cases the memory is either a read only memory
(ROM) or programmable read only memory (PROM). This contains a vast number of
points on the waveform, very many more than are accessed each cycle. A very large
number of points is required so that the phase accumulator can increment by a certain
number of points to set the required frequency.
The next stage in the process is to convert the digital numbers coming from the sine look
up table into an analogue voltage. This is achieved using a digital to analogue converter
(DAC). This signal is filtered to remove any unwanted signals and amplified to give the
required level as necessary.
Tuning is accomplished by increasing or decreasing the size of the step or phase
increment between different sample points. A larger increment at each update to the
phase accumulator will mean that the phase reaches the full cycle value faster and the
frequency is correspondingly high. Smaller increments to the phase accumulator value
means that it takes longer to increase the full cycle value and a correspondingly low value
of frequency. In this way it is possible to control the frequency. It can also be seen that
frequency changes can be made instantly by simply changing the increment value. There
is no need to a settling time as in the case of phase locked loop based synthesizer.
From this it can be seen that there is a finite difference between one frequency and the
next, and that the minimum frequency difference or frequency resolution is determined
by the total number of points available in the phase accumulator. A 24 bit phase
accumulator provides just over 16 million points and gives a frequency resolution of
about 0.25 Hz when used with a 5 MHz clock. This is more than adequate for most
purposes.
These synthesizers do have some disadvantages. There are a number of spurious signals
which are generated by a direct digital synthesizer. The most important of these is one
called an alias signal. Here images of the signal are generated on either side of the clock
frequency and its multiples. For example if the required signal had a frequency of 3 MHz
and the clock was at 10 MHz then alias signals would appear at 7 MHz and 13 MHz as

well as 17 MHz and 23 MHz etc.. These can be removed by the use of a low pass filter.
Also some low level spurious signals are produced close in to the required signal. These
are normally acceptable in level, although for some applications they can cause problems.
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Graphical method for designing a PLL frequency
synthesizer to meet a phase noise specification
- a simple graphical and understandable approach to understanding where
phase noise is generated within a PLL frequency synthesizer and designing
it to meet a requirement
Phase noise in PLL frequency synthesizers if of great importance because it determines
many factors about the equipment into which it is incorporated. For receivers it
determines the reciprocal mixing performance, and in some circumstances the bit error
rate. In transmitters the phase noise performance of the frequency synthesizer determines
features such as adjacent channel noise and it contributes to the bit error rate for the
whole system.
Phase noise in a synthesizer loop
Phase noise is generated at different points around the synthesizer loop and depending
upon where it is generated it affects the output in different ways. For example, noise
generated by the VCO has a different effect to that generated by the phase detector. This
illustrates that it is necessary to look at the noise performance of each circuit block in the
loop when designing the synthesizer so that the best noise performance is obtained.
Apart from ensuring that the noise from each part of the circuit is reduced to an absolute
minimum, it is the loop filter which has the most effect on the final performance of the
circuit because it determines the break frequencies where noise from different parts of the
circuit start to affect the output.
To see how this happens take the example of noise from the VCO. Noise from the
oscillator is divided by the divider chain and appears at the phase detector. Here it
appears as small perturbations in the phase of the signal and emerges at the output of the
phase detector. When it comes to the loop filter only those frequencies which are below
its cut-off point appear at the control terminal of the VCO to correct or eliminate the
noise. From this it can be seen that VCO noise which is within the loop bandwidth is
attenuated, but that which is outside the loop bandwidth is left unchanged.
The situation is slightly different for noise generated by the reference. This enters the
phase detector and again passes through it to the loop filter where the components below
the cut-off frequency are allowed through and appear on the control terminal of the VCO.
Here they add noise to the output signal. So it can be seen that noise from the reference is
added to the output signal within the loop bandwidth but it is attenuated outside this.
Similar arguments can be applied to all the other circuit blocks within the loop. In
practice the only other block which normally has any major effect is the phase detector
and its noise affects the loop in exactly the same way as noise from the reference. Also if
multi-loop synthesizers are used then the same arguments can be used again.
Effects of multiplication
As noise is generated at different points around the loop it is necessary to discover what
effect this has on the output. As a result it is necessary to relate all the effects back to the
VCO. Apart from the different elements in the loop affecting the noise at the output in
different ways, the effect of the multiplication in the loop also has an effect.
The effect of multiplication is very important. It is found that the level of phase noise
from some areas is increased in line with the multiplication factor (i.e. the ratio of the

final output frequency to the phase comparison frequency). In fact it is increased by a
factor of 20 log10 N where N is the multiplication factor. The VCO is unaffected by this,
but any noise from the reference and phase detector undergoes this amount of
degradation. Even very good reference signals can be a major source of noise if the
multiplication factor is high. For example a loop which has a divider set to 200 will
multiply the noise of the reference and phase detector by 46 dB.
From this information it is possible to build up a picture of the performance of the
synthesizer. Generally this will look like the outline shown in Fig. 6. From this it can be
seen that the noise inside the loop bandwidth is due mainly to components like the phase
detector and reference, whilst outside the loop the VCO generates the noise. A slight
hump is generally seen at the point where the loop filter cuts off and the loop gain falls to
unity.
By predicting the performance of the loop it is possible to optimise the performance or
look at areas which can be addressed to improve the performance of the whole
synthesizer before the loop is even built. In order to analyse the loop further it is
necessary to look at each circuit block in turn.
Voltage controlled oscillator
The noise performance of the oscillator is of particular importance. This is because the
noise performance of the synthesizer outside the loop is totally governed by its
performance. In addition to this its performance may influence decisions about other
areas of the circuit.
The typical noise outline for a VCO is flat at large frequency offsets from the carrier. It is
determined largely by factors such as the noise figure of the active device. The
performance of this area of the oscillator operation can be optimised by ensuring the
circuit is running under the optimum noise performance conditions. Another approach is
to increase the power level of the circuit so that the signal to noise ratio improves.
Closer in the noise starts to rise, initially at a rate of 20 dB per decade. The point at which
this starts to rise is determined mainly by the Q of the oscillator circuit. A high Q circuit
will ensure a good noise performance. Unfortunately VCOs have an inherently low Q
because of the Q of the tuning varactors normally employed. Performance can be
improved by increasing the Q, but this often results in the coverage of the oscillator being
reduced.
Still further in towards the carrier the noise level starts to rise even faster at a rate of 30
dB per decade. This results from flicker or 1/f noise. This can be improved by increasing
the level of low frequency feedback in the oscillator circuit. In a standard bipolar circuit a
small un-bypassed resistor in the emitter circuit can give significant improvements.
To be able to assess the performance of the whole loop it is necessary to assess the
performance of the oscillator once it has been designed and optimised. Whilst there are a
number of methods of achieving this the most successful is generally to place the
oscillator into a loop having a narrow bandwidth and then measure its performance with a
spectrum analyser. By holding the oscillator steady this can be achieved relatively easily.
However the results are only valid outside the loop bandwidth. However a test loop is
likely to have a much narrower bandwidth than the loop being designed the noise levels
in the area of interest will be unaltered.
Reference
The noise performance of the reference follows the same outlines as those for the VCO,
but the performance is naturally far better. The reason for this is that the Q of the crystal
is many orders of magnitude higher than that of the tuned circuit in the VCO.
Typically it is possible to achieve figures of -110 dBc/Hz at 10 Hz from the carrier and
140 dBc/Hz at 1 kHz from a crystal oven. Figures of this order are quite satisfactory for
most applications. If lower levels of reference noise are required these can be obtain, but
at a cost. In instances where large multiplication factors are necessary a low noise
reference may be the only option. However as a result of the cost they should be avoided

wherever possible. Plots of typical levels of phase noise are often available with crystal
ovens giving an accurate guide to the level of phase noise generated by the reference.
Frequency divider
Divider noise appears within the loop bandwidth. Fortunately the levels of noise
generated within the divider are normally quite low. If an analysis is required then it will
be found that noise is generated at different points within the divider each of which will
be subject to a different multiplication factor dependent upon where in the divider it is
generated and the division ratio employed from that point.
Most divider chains use several dividers and if an approximate analysis is to be
performed it may be more convenient to only consider the last device or devices in the
chain as these will contribute most to the noise. However the noise is generally difficult
to measure and will be seen with that generated by the phase detector.
Phase detector
Like the reference signal the phase detector performance is crucial in determining the
noise performance within the loop bandwidth. There are a number of different types of
phase detector. The two main categories are analogue and digital.
Mixers are used to give analogue phase detectors. If the output signal to noise ratio is to
be as good as possible then it is necessary to ensure that the input signal levels are as high
as possible within the operating limits of the mixer. Typically the signal input may be
limited to around -10 dBM and the local oscillator input to +10 dBm. In some instances
higher level mixers may be used with local oscillator levels of +17 dBm or higher. The
mixer should also be chosen to have a low NTR (noise temperature ratio). As the output
is DC coupled it is necessary to have a low output load resistance to prevent a backward
bias developing. This could offset the operation of the mixer and reduce its noise
performance.
It is possible to calculate the theoretical noise performance of the mixer under optimum
conditions. An analogue mixer is likely to give a noise level of around -153 dBc/Hz.
There are a variety of digital phase detectors which can be used. In theory these give a
better noise performance than the analogue counterpart. At best a simple OR gate type
will give figures about 10 dB better than an analogue detector and an edge triggered type
(e.g. a dual D type or similar) will give a performance of around 5 dB better than the
analogue detector.
These figures are the theoretical optimum and should be treated as guide although they
are sufficient for initial noise estimates. In practice other factors may mean that the
figures are different. A variety of factors including power supply noise, circuit layout etc.
can degrade the performance from the ideal. If very accurate measurements are required
then results from the previous use of the circuit, or from a special test loop can provide
the required results.
Loop filter
There are a variety of parameters within the area of the loop filter which affect the noise
performance of the loop. The break points of the filter and the unity gain point of the loop
determined by the filter govern the noise profile.
In terms of contributions to the noise in the loop the major source is likely to occur if an
operational amplifier is used. If this is the case a low noise variety must be used
otherwise the filter will give a large contribution to the loop phase noise profile. This
noise is often viewed as combined with that from the phase detector, appearing to
degrade its performance from the ideal.
Plotting Performance
Having investigated the noise components from each element in the loop, it is possible to
construct a picture of how the whole loop will perform. Whilst this can performed
mathematically, a simple graphical approach quickly reveals an estimate of the
performance and shows which are the major elements which contribute to the noise. In

this way some re-design can be undertaken before the design is constructed, enabling the
best option to be chosen on the drawing board. Naturally it is likely to need some
optimisation once it has been built, but this method enables the design to be made as
close as possible beforehand.
First it is necessary to obtain the loop response. This is dependent upon a variety of
factors including the gain around the loop and the loop filter response. For stability the
loop gain must fall at a rate of 20 dB per decade (6 dB per octave) at the unity gain point.
Provided this criterion is met a wide variety of filters can be used. Often it is useful to
have the loop response rise at a greater rate than this inside the loop bandwidth. By doing
this the VCO noise can be attenuated further. Outside the loop bandwidth a greater fall
off rate can aid suppress the unwanted reference sidebands further. From a knowledge of
the loop filter chosen the break points can be calculated and with a knowledge of the loop
gain the total loop response can be plotted.
With the response known the components from the individual blocks in the loop can be
added as they will be affected by the loop and seen at the output.
First take the VCO. Outside the loop bandwidth its noise characteristic is unmodified.
However once inside this point the action of the loop attenuates the noise, first at a rate of
20 dB per decade, and then at a rate of 40 dB per decade. The overall affect of this is to
modify the response of the characteristic as shown in Fig. 10. It is seen that outside the
loop bandwidth the noise profile is left unmodified. Far out the noise is flat, but further in
the VCO noise rises at the rate of 20 dB per decade. Inside the loop bandwidth the VCO
noise will be attenuated first at the rate of 20 dB per decade, which in this case gives a
flat noise profile. Then as the loop gain increases at the filter break point, to 40 dB per
decade this gives a fall in the VCO noise profile of -20 dB per decade. However further
in the profile of the stand alone VCO noise rises to -30dB per decade. The action of the
loop gives an overall fall of -10 dB per decade.
The effects of the other significant contributions can be calculated. The reference
response can easily be deduced from the manufacturers figures. Once obtained these must
have the effect of the loop multiplication factor added. Once this has been calculated the
effect of the loop can be added. Inside the loop there is no effect on the noise
characteristic, however outside this frequency it will attenuate the reference noise, first at
a rate of 20 dB per decade and then after the filter break point at 40 dB per decade.
The other major contributor to the loop noise is the phase detector. The effect of this is
treated in the same way as the reference, having the effect of the loop multiplication
added and then being attenuated outside the loop bandwidth.
Once all the individual curves have been generated they can be combined onto a single
plot to gain a full picture of the performance of the synthesizer. When doing this it should
be remembered that it is necessary to produce the RMS sum of the components because
the noise sources are not correlated.
Once this has been done then it is possible to optimise the performance by changing
factors like the loop bandwidth, multiplication factor and possibly the loop topology to
obtain the best performance and ensure that the required specifications are met. In most
cases the loop bandwidth is chosen so that a relatively smooth transition is made between
the noise contributions inside and outside the loop. This normally corresponds to lowest
overall noise situation.
Summary
Although this approach may appear to be slightly "low tech" in today's highly
computerised engineering environment it has the advantage that a visual plot of the
predicted performance can be easily put together. In this way the problem areas can be
quickly identified, and the noise performance of the whole synthesizer optimised before
the final design is committed.
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Radio receiver amplitude modulation (AM)
demodulation
- using a simple diode detector ( demodulator )
One of the advantages of amplitude modulation (AM) is that it is cheap and easy to build
a demodulator circuit for a radio receiver. The simplicity AM radio receivers AM is one
of the reasons why AM has remained in service for broadcasting for so long. One of the
key factors of this is the simplicity of the receiver AM demodulator.
A number of methods can be used to demodulate AM, but the simplest is a diode
detector. It operates by detecting the envelope of the incoming signal. It achieves this by
simply rectifying the signal. Current is allowed to flow through the diode in only one
direction, giving either the positive or negative half of the envelope at the output. If the
detector is to be used only for detection it does not matter which half of the envelope is
used, either will work equally well. Only when the detector is also used to supply the
automatic gain control (AGC) circuitry will the polarity of the diode matter.
The AM detector or demodulator includes a capacitor at the output. Its purpose is to
remove any radio frequency components of the signal at the output. The value is chosen
so that it does not affect the audio base-band signal. There is also a leakage path to enable
the capacitor to discharge, but this may be provided by the circuit into which the
demodulator is connected.
A simple diode detector (demodulator) for AM signals
This type of detector or demodulator is called a linear envelope detector because the
output is proportional to the input envelope. Unfortunately the diodes used can introduce
appreciable levels of harmonic distortion unless modulation levels are kept low. As a
result these detectors can never provide a signal suitable for high quality applications.
Additionally these detectors ( demodulators ) are susceptible to the effects of selective
fading experienced on short wave broadcast transmissions. Here the ionospheric
propagation may be such that certain small bands of the signal are removed. Under
normal circumstances signals received via the ionosphere reach the receiver via a number
of different paths. The overall signal is a combination of the signals received via each
path and as a result they will combine with each other, sometimes constructively to
increase the overall signal level and sometimes destructively to reduce it. It is found that
when the path lengths are considerably different this combination process can mean that
small portions of the signal are reduced in strength. An AM signal consists of a carrier
with two sidebands.

Spectrum of an amplitude modulated (AM) signal
If the section of the signal that is removed falls in one of the sidebands, it will change the
tone of the received signal. However if carrier is removed or even reduced in strength, the
signal will appear to be over modulated, and severe distortion will result. This is a
comparatively common occurrence on the short waves, and means that diode detectors
are not suitable for high quality reception. Synchronous demodulation ( detection ) is far
superior.
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Synchronous demodulation / detection
Today's radio receivers offer very high levels of performance and boast many facilities.
Many radio receivers incorporate memories, phase locked loops, direct digital synthesis,
digital signal processing and much more. One facility that can be very useful on the short
wave bands is synchronous detection or synchronous demodulation as this can give much
improved performance for receiving amplitude modulation (AM) transmissions.
Unfortunately little is written about this form of modulation, and often it is a matter of
accepting that it must be better than any normal options because it is included as a feature
in the receiver specification.
Synchronous detection is used for the detection or demodulation of amplitude modulation
(AM). This form of modulation is still widely used for broadcasting on the long, medium
and short wave bands despite the fact that there are more efficient forms of modulation
that can be used today. The main reason for its use nowadays is that it is very well
established, and there are many millions of AM receivers around the world today.
In any receiver a key element is the detector. Its purpose is to remove the modulation
from the carrier to give the audio frequency representation of the signal. This can be
amplified by the audio amplifier ready to be converted into audible sound by headphones
or a loudspeaker. Many receivers still use what is termed an envelope detector using a
semiconductor diode for demodulating AM. These detectors have a number of
disadvantages. The main one is that they are not particularly linear and distortion levels
may be high. Additionally their noise performance is not particularly good at low signal
levels.
These detectors also do not perform very well when the signal undergoes selective fading
as often occurs on the short wave bands. An AM signal contains two sidebands and the
carrier. For the signal to be demodulated correctly the carrier should be present at the
required level. It can be seen that the signal covers a definite bandwidth, and the effects
of fading may result in the carrier and possibly one of the sidebands being reduced in
level. If this occurs then the received signal appears to be over-modulated with the result
that distortion occurs in the demodulation process.

The spectrum of an amplitude modulated signal
Diode envelope detector
In virtually every receiver a simple diode envelope detector is used. These circuits have
the advantage that they are very simple and give adequate performance in many
applications.
The circuit of a typical detector is shown in Figure 2. Here the diode first rectifies the
signal to leave only the positive or negative going side of the signal, and then a capacitor
removes any of the remaining radio frequency components to leave the demodulated
audio signal. Unfortunately diodes are not totally linear and this is the cause of the
distortion.
An envelope detector for AM signals
What is synchronous demodulation
Signals can be demodulated using a system known as synchronous detection or
demodulation. This is far superior to diode or envelope detection, but requires more
circuitry. Here a signal on exactly the same frequency as the carrier is mixed with the
incoming signal as shown in Figure 2. This has the effect of converting the frequency of
the signal directly down to audio frequencies where the sidebands appear as the required
audio signals in the audio frequency band.
The crucial part of the synchronous detector is in the production a local oscillator signal
on exactly the same frequency as the carrier. Although it is possible to receive an AM
signal without the local oscillator frequency on exactly the same frequency as the carrier
this is the same as using the BFO in a receiver to resolve the signal. If the BFO is not
exactly on the same frequency as the carrier then the resultant audio is not very good.
Synchronous demodulation

Fortunately this is not too difficult to achieve and although there are a number of ways of
achieving this the most commonly used method is to pass some of the signal into a high
gain limiting amplifier. The gain of the amplifier is such that it limits, and thereby
removing all the modulation. This leaves a signal consisting only of the carrier and this
can be used as the local oscillator signal in the mixer as shown in Fig. 4. This is most
convenient, cheapest and certainly the most elegant method of producing synchronous
demodulation.
A synchronous detector using a high gain-limiting amplifier to extract the carrier
Advantages of synchronous detection
A synchronous detector is more expensive to make than an ordinary diode detector when
discrete components are used, although with integrated circuits being found in many
receivers today there is little or no noticeable cost associated with its use as the circuitry
is often included as part of an overall receiver IC.
Synchronous detectors are used because they have several advantages over ordinary
diode detectors. Firstly the level of distortion is less. This can be an advantage if a better
level of quality is required but for many communications receivers this might not be a
problem. Instead the main advantages lie in their ability to improve reception under
adverse conditions, especially when selective fading occurs or when signal levels are low.
Under conditions when the carrier level is reduced by selective fading, the receiver is
able to re-insert its own signal on the carrier frequency ensuring that the effects of
selective fading are removed. As a result the effects of selective fading can be removed to
greatly enhance reception.
The other advantage is an improved signal to noise ratio at low signal levels. As the
demodulator is what is termed a coherent modulator it only sees the components of noise
that are in phase with the local oscillator. Consequently the noise level is reduced and the
signal to noise ratio is improved.
Unfortunately synchronous detectors are only used in a limited number of receivers
because of their increased complexity. Where they are used a noticeable improvement in
receiver performance is seen and when choosing a receiver that will be used for short
wave broadcast reception it is worth considering whether a synchronous detector is one
of the facilities that is required.
Frequency modulation is widely used in radio communications and broadcasting,
particularly on frequencies above 30 MHz. It offers many advantages, particularly in
mobile radio applications where its resistance to fading and interference is a great
advantage. It is also widely used for broadcasting on VHF frequencies where it is able to
provide a medium for high quality audio transmissions.
In view of its widespread use receivers need to be able to demodulate these
transmissions. There is a wide variety of different techniques and circuits that can be sued
including the Foster-Seeley, and ratio detectors using discreet components, and where
integrated circuits are used the phase locked loop and quadrature detectors are more
widely used.

What is FM?
As the name suggests frequency modulation uses changes in frequency to carry the sound
or other information that is required to be placed onto the carrier. As shown in Figure 1 it
can be seen that as the modulating or base band signal voltage varies, so the frequency of
the signal changes in line with it. This type of modulation brings several advantages with
it. The first is associated with interference reduction. Much interference appears in the
form of amplitude variations and it is quite easy to make FM receivers insensitive to
amplitude variations and accordingly this brings about a reduction in the levels of
interference. In a similar way fading and other strength variations in the signal have little
effect. This can be particularly useful for mobile applications where charges in location
as the vehicle moves can bring about significant signal strength changes. A further
advantage of FM is that the RF amplifiers in transmitters do not need to be linear. When
using amplitude modulation or its derivatives, any amplifier after the modulator must be
linear otherwise distortion is introduced. For FM more efficient class C amplifiers may be
used as the level of the signal remains constant and only the frequency varies.
Frequency modulating a signal
Wide band and Narrow band
When a signal is frequency modulated, the carrier shifts in frequency in line with the
modulation. This is called the deviation. In the same way that the modulation level can be
varied for an amplitude modulated signal, the same is true for a frequency modulated one,
although there is not a maximum or 100% modulation level as in the case of AM.
The level of modulation is governed by a number of factors. The bandwidth that is
available is one. It is also found that signals with a large deviation are able to support
higher quality transmissions although they naturally occupy a greater bandwidth. As a
result of these conflicting requirements different levels of deviation are used according to
the application that is used.
Those with low levels of deviation are called narrow band frequency modulation
(NBFM) and typically levels of +/- 3 kHz or more are used dependent upon the
bandwidth available. Generally NBFM is used for point to point communications. Much
higher levels of deviation are used for broadcasting. This is called wide band FM
(WBFM) and for broadcasting deviation of +/- 75 kHz is used.
Receiving FM
In order to be able to receive FM a receiver must be sensitive to the frequency variations
of the incoming signals. As already mentioned these may be wide or narrow band.
However the set is made insensitive to the amplitude variations. This is achieved by
having a high gain IF amplifier. Here the signals are amplified to such a degree that the
amplifier runs into limiting. In this way any amplitude variations are removed.

In order to be able to convert the frequency variations into voltage variations, the
demodulator must be frequency dependent. The ideal response is a perfectly linear
voltage to frequency characteristic. Here it can be seen that the centre frequency is in the
middle of the response curve and this is where the un-modulated carrier would be located
when the receiver is correctly tuned into the signal. In other words there would be no
offset DC voltage present.
The ideal response is not achievable because all systems have a finite bandwidth and as a
result a response curve known as an "S" curve is obtained. Outside the badwidth of the
system, the response falls, as would be expected. It can be seen that the frequency
variations of the signal are converted into voltage variations which can be amplified by
an audio amplifier before being passed into headphones, a loudspeaker, or passed into
other electronic circuitry for the appropriate processing.
Characteristic "S" curve of an FM demodulator
To enable the best detection to take place the signal should be centred about the middle of
the curve. If it moves off too far then the characteristic becomes less linear and higher
levels of distortion result. Often the linear region is designed to extend well beyond the
bandwidth of a signal so that this does not occur. In this way the optimum linearity is
achieved. Typically the bandwidth of a circuit for receiving VHF FM broadcasts may be
about 1 MHz whereas the signal is only 200 kHz wide.
Demodulators
There are a number of circuits that can be used to demodulate FM. Each type has its own
advantages and disadvantages, some being used when receivers used discrete
components, and others now that ICs are widely used.
Slope detection
The very simplest form of FM demodulation is known as slope detection or
demodulation. It simply uses a tuned circuit that is tuned to a frequency slightly offset
from the carrier of the signal. As the frequency of the signal varies up and down in
frequency according to its modulation, so the signal moves up and down the slope of the
tuned circuit. This causes the amplitude of the signal to vary in line with the frequency
variations. In fact at this point the signal has both frequency and amplitude variations.
The final stage in the process is to demodulate the amplitude modulation and this can be
achieved using a simple diode circuit. One of the most obvious disadvantages of this
simple approach is the fact that both amplitude and frequency variations in the incoming
signal appear at the output. However the amplitude variations can be removed by placing
a limiter before the detector. Additionally the circuit is not particularly efficient as it
operates down the slope of the tuned circuit. It is also unlikely to be particularly linear,
especially if it is operated close to the resonant point to minimise the signal loss.
Ratio and Foster-Seeley detectors
When circuits employing discrete components were more widely sued, the Ratio and
Foster-Seeley detectors were widely used. Of these the ratio detector was the most
popular as it offers a better level of amplitude modulation rejection of amplitude

modulation. This enables it to provide a greater level of noise immunity as most noise is
amplitude noise, and it also enables the circuit to operate satisfactorily with lower levels
of limiting in the preceding IF stages of the receiver.
The operation of the ratio detector centres around a frequency sensitive phase shift
network with a transformer and the diodes that are effectively in series with one another.
When a steady carrier is applied to the circuit the diodes act to produce a steady voltage
across the resistors R1 and R2, and the capacitor C3 charges up as a result.
The transformer enables the circuit to detect changes in the frequency of the incoming
signal. It has three windings. The primary and secondary act in the normal way to
produce a signal at the output. The third winding is un-tuned and the coupling between
the primary and the third winding is very tight, and this means that the phasing between
signals in these two windings is the same.
The primary and secondary windings are tuned and lightly coupled. This means that there
is a phase difference of 90 degrees between the signals in these windings at the centre
frequency. If the signal moves away from the centre frequency the phase difference will
change. In turn the phase difference between the secondary and third windings also
varies. When this occurs the voltage will subtract from one side of the secondary and add
to the other causing an imbalance across the resistors R1 and R2. As a result this causes a
current to flow in the third winding and the modulation to appear at the output.
The capacitors C1 and C2 filter any remaining RF signal which may appear across the
resistors. The capacitor C4 and R3 also act as filters ensuring no RF reaches the audio
section of the receiver.
The ratio detector
The Foster Seeley detector has many similarities to the ratio detector. The circuit
topology looks very similar, having a transformer and a pair of diodes, but there is no
third winding and instead a choke is used.

The Foster-Seeley detector
Like the ratio detector, the Foster-Seeley circuit operates using a phase difference
between signals. To obtain the different phased signals a connection is made to the
primary side of the transformer using a capacitor, and this is taken to the centre tap of the
transformer. This gives a signal that is 90 degrees out of phase.
When an un-modulated carrier is applied at the centre frequency, both diodes conduct, to
produce equal and opposite voltages across their respective load resistors. These voltages
cancel each one another out at the output so that no voltage is present. As the carrier
moves off to one side of the centre frequency the balance condition is destroyed, and one
diode conducts more than the other. This results in the voltage across one of the resistors
being larger than the other, and a resulting voltage at the output corresponding to the
modulation on the incoming signal.
The choke is required in the circuit to ensure that no RF signals appear at the output. The
capacitors C1 and C2 provide a similar filtering function.
Both the ratio and Foster-Seeley detectors are expensive to manufacture. Wound
components like coils are not easy to produce to the required specification and therefore
they are comparatively costly. Accordingly these circuits are rarely used in modern
equipment.
Quadrature detector
Another form of FM detector or demodulator that can be these days is called the
quadrature detector. It lends itself to use with integrated circuits and as a result it is in
widespread use. It has the advantage over the ratio and Foster-Seeley detectors that it
only requires a simple tuned circuit.
For the quadrature detector, the signal is split into two components. One passes through a
network that provides a basic 90 degree phase shift, plus an element of phase shift
dependent upon the deviation and into one port of a mixer. The other is passed straight
into another port of the mixer. The output from the mixer is proportional to the phase
difference between the two signals, i.e. it acts as a phase detector and produces a voltage
output that is proportional to the phase difference and hence to the level of deviation on
the signal.
The detector is able to operate with relatively low input levels, typically down to levels of
around 100 microvolts and it is very easy to set up requiring only the phase shift network
to be tuned to the centre frequency of the expected signal. It also provides good linearity
enabling very low levels of distortion to be achieved.
Often the analogue multiplier is replaced by a logic AND gate. The input signal is hard
limited to produce a variable frequency pulse waveform. The operation of the circuit is
fundamentally the same, but it is known as a coincidence detector. Also the output of the

AND gate has an integrator to "average" the output waveform to provide the required
audio output, otherwise it would consist of a series of square wave pulses.
Phase locked loop (PLL)
Another popular form of FM demodulator comes in the form of a phase locked loop. Like
the quadrature detector, phase locked loops do not need to use a coil, and therefore they
make a very cost effective form of demodulator.
The way in which they operate is very simple. The loop consists of a phase detector into
which the incoming signal is passed, along with the output from the voltage controlled
oscillator (VCO) contained within the phase locked loop. The output from the phase
detector is passed into a loop filter and then sued as the control voltage for the VCO.
Phase locked loop (PLL) FM demodulator
With no modulation applied and the carrier in the centre position of the pass-band the
voltage on the tune line to the VCO is set to the mid position. However if the carrier
deviates in frequency, the loop will try to keep the loop in lock. For this to happen the
VCO frequency must follow the incoming signal, and for this to occur the tune line
voltage must vary. Monitoring the tune line shows that the variations in voltage
correspond to the modulation applied to the signal. By amplifying the variations in
voltage on the tune line it is possible to generate the demodulated signal.
It is found that the linearity of this type of detector is governed by the voltage to
frequency characteristic of the VCO. As it normally only swings over a small portion of
its bandwidth, and the characteristic can be made relatively linear, the distortion levels
from phase locked loop demodulators are normally very low.
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Radio receiver filter options
- including LC filter, crystal filter, mechanical filter, ceramic filter, and roofing
filter
There is a wide variety of different types of filter used within superhet radios. Some
radios will simply use filters made up from the tuned transformers (LC filters based on
capacitors and inductors) linking the different intermediate frequency stages within the
radios or used with an IC in the radio. Other radio receivers may incorporate highly
selective crystal filters, whereas others may use mechanical filters (like those used by the
Collins Radio Company some years ago) or ceramic filters. Each radio will have its own
requirements, and the choice of filter to meet its needs of performance and cost.
LC tuned circuits
The simplest type of filter is an ordinary L-C tuned circuit. In many older radio using
discrete semiconductors, or older radios using vacuum tubes they take the form of
transformers to couple the individual stages in an IF amplifier chain. Often there are two
or three stages with tuned circuits. Using them it is usually possible to achieve sufficient
selectivity for a medium wave AM or VHF FM broadcast radio. However for a good
quality communications receiver it is rarely possible to be able to achieve the required
degree of selectivity using just L-C filters.

In more modern radios using integrated circuits a single tuned circuit could be used in
conjunction with an integrated, as the concept of inter-stage coupling is not employed in
the same manner. Typically a ceramic filter, rather than an LC circuit is more likely to be
used.
If L-C filters were used in a radio using interstage transformers then it would be possible
to increase the degree of selectivity by increasing the number of tuned circuits between
each stage. This is not ideal for a number of reasons. In the first case it increases the
difficulty of aligning the set. In addition to this each tuned circuit will introduce a certain
amount of loss. Increasing the number of tuned circuits will increase the amount of gain
required, sometimes necessitating a further stage of gain. A further disadvantage is that it
is not easy to alter the degree of selectivity by switching in additional L-C filters. If this is
to be achieved then it is often preferable to switch in a further type of filter such as a
crystal filter.
Crystal Filters
Crystal filters provide the main selectivity in of most of today's high performance sets.
They provided exceedingly high degrees of selectivity which are hard to equal in terms of
performance and cost.
The crystals in the filters are made from a substance called quartz. This is basically a
form of crystalline silicon. Originally natural deposits were used to manufacture the
crystals required for the electronics industry. Now quartz crystals are grown synthetically
under controlled conditions to produce very high quality material.
The crystals use the piezo-electric effect for their operation. This effect occurs in a
number of substances and it converts a mechanical stress into a voltage and vice versa.
Many electrical transducers use the effect converting electrical impulses or signals into
mechnical vibrations and vice versa.
In quartz crystal resonators the piezo-electric effect is used in conjunction with the
mechanical resonances which occur in the substance. The electrical signals passing into
the crystal are converted into mechanical vibrations which interact with the resonances of
the crystal. In this way the crystal uses the piezo-electric effect to enable the mechanical
resonances to tune the electrical signals. These mechanical resonances have exceedingly
high Q factors. Many crystals will exhibit values of several thousand. This is many orders
of magnitude higher than ordinary tuned filters made from conventional inductors and
capacitors where values of a hundred or so are considered high. Typically the Q of an LC
tuned circuit may be reach values of a few hundred. For quartz crystals values of Q may
exceed 100 000.
Further details about quartz, its properties and the ways in which crystals are
manufactured and used can be found on the Electronic components section of this site -
see side menu for the link.
The response of a single crystal is too narrow for many applications. Normally a filter is
required to have a passband, possibly of a few hundred Hertz, or a few kilohertz, and
outside this bandwidth, other signals should be totally rejected. While it is not possible to
achieve the perfect filter very high degrees of selectivity can be achieved. By adding
several crystals together it is possible to obtain the performance that is required. Often
crystal filters are referred to as having a certain number of poles. This terminology comes
from the filter analysis design process, but effectively there is one crystal in the filter for
every pole.
A two pole filter (i.e. one with two crystals) is not normally adequate to meet many
requirements. The shape factor which is the ratio between the bandwidth where the
stopband attenuation starts and the bandwidth of the passband) can be greatly improved
by adding further sections. Typically ultimate rejections of 70 dB and more are required
in a receiver. As a rough guide a two pole filter will generally give a rejection of around
20 dB; a four pole filter, 50 dB; a six pole filter, 70 dB; and an eight pole one 90 dB.

Monolithic filters
With more items being integrated onto single chips these days it is hardly surprising to
find that a similar approach is being adopted for crystal filters. Instead of having several
separate or discrete crystals in a filter, even if they are all contained in the same can, it is
possible to put a complete filter onto a single quartz crystal, hence the name monolithic
crystal filter.
In essence the filter is made up by placing two sets of electrodes at opposite sides of a
single AT cut crystal. The coupling between the two electrodes acts in such a way that a
highly selective filter is produced.
Monolithic filters have only been available since the 1970s. Even now a large number of
filter manufacturers do not produce them, preferring to use the more traditional filters
made from individual crystals.
While it had been known for a long while that a two pole filter could be made up on a
single crystal, the idea was not developed because the way in which it worked was not
understood. After much work, scientists at Bell Laboratories in the USA discovered its
mode of operation. Very simply it consists of two acoustically coupled resonators.
A monolithic crystal filter consists of a crystal blank onto which two sets of electrodes or
plates are placed at opposite ends of the blank. Each set consists of an electrode on either
side of the blank. When the electrical signal is placed across one pair of electrodes, the
piezo-electric effect converts this into mechanical vibrations. These travel across the
crystal to the other electrodes where they are converted back into an electrical signal
again. However if the acoustic signal is to travel across the crystal then its frequency
must match the resonance of the crystal.
Often these filters are manufactured for operation below about 30 MHz, because above
these frequencies the manufacturing costs tend to rise. However manufacturing
techniques are improving all the time it is possible to use them above this. If this is
required then the normal way of accomplishing this is to use an overtone mode. This
considerably increases the maximum possible frequencies, although the performance is
not usually quite as good.
Monolithic filters are used in many areas now. They offer better performance than their
discrete counterparts and they can be made smaller - a feature which is becoming
increasingly important in today's miniaturised electronics industry. The main drawback of
these filters is that they require very specialised equipment for their manufacture.
Ceramic filters
Quartz is not the only substance to exhibit the piezo-electric effect combined with a sharp
resonance. A number of ceramics are also used successfully to perform this function.
Although filters made from these ceramics are not nearly as selective as their higher
quality quartz relatives, they are cheaper and offer great improvements over their L-C
counterparts.
Ceramic filters are made from a specialised family of ceramics, and the elements for
filters are normally in the form of a small disc. They operate in exactly the same way as
crystal filters, the signal being linked to the mechanical resonances by the piezo-electric
effect. Generally ceramic filters have a much wider bandwidth and a poorer shape factor
than their crystal counterparts. As a result they are rarely used in high performance
communications receivers as the primary form of filtering, although their performance
has improved dramatically in recent years and some examples of ceramic filters offering
exceedingly good levels of performance are available. As a result they find widespread
use in broadcast receivers for AM and VHF FM reception and some wireless
applications.
Mechanical filters
When high performance filters are needed there is another type which can be considered.
Although not nearly as popular as crystal filters these days, mechanical filters found
widespread use a number of years ago. The Collins Radio Company (now Rockwell

Collins) was a famous manufacturer of these devices, introducing their first designs in
1952, these filters are still manufactured.
In essence their operation is very similar to that of a crystal, although the various
functions are performed by individual components within the filter. At either end of the
filter assembly there are transducers which convert the signals from their electrical form
to mechanical vibrations, and back again at the other end. These vibrations are applied to
a series of discs which are mechanically resonant at the required frequency. Each of these
discs has a Q of which can be about 5000 or more, and they are arranged close to one
another but not touching to form a long cylinder. A number of coupling rods are attached
to run along the side of the assembly to transfer the vibrations from one section to the
next. By altering the amount of coupling between the sections and the resonance of each
disc, the response of the overall unit can be tailored to meet the exact requirements.
Operation of these mechanical filters is normally confined to frequencies between about
50 and 500 kHz. Below these frequencies the discs become too large, whilst at the top
end of the range they are too small to manufacture and mount in the filters with any
degree of reliability. Apart from the limited frequency range the other disadvantage is
that the resonant frequency of these filters drifts with temperature. However one of their
main advantages is that exceedingly narrow bandwidths can be achieved relatively easily,
and the low levels of intermodulation distortion they introduce. Additionally the costs of
these devices have been reduced over the years and the number of resonators that can be
used can be between 2 and 12 dependent upon the requirements.
Roofing filters
In many receivers the main filtering occurs only after there have been many stages of
amplification. This means that a strong signal which is outside the pass-band of the main
receiver filter can cause overloading especially in the early IF stages before the filter.
This occurs because the AGC does not see the signal and reduce the gain of the earlier
stages to take account of it, or the operator may not be aware of the signal and reduce the
RF gain if a control is available.
To overcome this problem a wider bandwidth filter is placed early on in the IF stages to
reduce the level of any strong off channel signals. The main filtering, however, is still
provided late on in the receiver by the main full specification filter.
Roofing filters are often found in multi-conversion superhet receivers where the main
filter is found after two or possibly three conversion stages. The roofing filter can be
placed soon after the first mixer to reduce the effects of any strong off-channel signals.
Summary
There is a good selection of filters that can be used in radio receivers. The actual type that
is eventually decided upon a balance of performance, cost and other factors. For many
applications where the highest levels of performance are not needed, ceramic filters
provide the ideal solution being very cheap and easy to use while providing levels of
performance that are quite adequate for many applications. For applications where only
the highest levels of performance are required, crystal filters are the most common
solution either as units made from discrete crystals or as monolithic filters. However
mechanical filters could ebb considered for some applications. These days LC filters are
not widely used because the cost of winding coils is high, and often ceramic filters are
more convenient, cheaper, and offer a better level of performance.
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Quartz crystal band pass filters
- including the single crystal filter, half lattice filter and ladder filter for use in radio
receivers

Crystal filters are widely used in many applications including radio receivers. The very
high level of Q makes them ideal for use as the primary band pass filter in a radio
receiver. As a result of this there are a number of circuits that have been used to provide
the required level of selectivity and performance over the years. These include the single
crystal filter, the half lattice crystal filter and the ladder filter.
Single crystal filter
The simplest crystal filter employs a single crystal. This type of filter was developed in
the 1930s and was used in early receivers dating from before the 1960s but is rarely used
today. Although it employs the very high Q of the crystal, its response is asymmetric and
it is too narrow for most applications, having a bandwidth of a hundred Hz or less.
In the circuit there is a variable capacitor that is used to compensate for the parasitic
capacitance in the crystal. This capacitor was normally included as a front panel control.
Diagram of filter using a single quartz crystal
Half lattice crystal filter
This form of band pass filter provided a significant improvement in performance over the
single. In this configuration the parasitic capacitances of each of the crystals cancel each
other out and enable the circuit to operate satisfactorily. By adopting a slightly different
frequency for the crystals, a wider bandwidth is obtained. However the slope response
outside the required pass band falls away quickly, enabling high levels of out of band
rejection to be obtained. Typically the parallel resonant frequency of one crystal is
designed to be equal to the series resonant frequency of the other.
Despite the fact that the half lattice crystal filter can offer a much flatter in-band response
there is still some ripple. This results from the fact that the two crystals have different
resonant frequencies. The response has a small peak at either side of the centre frequency
and a small dip in the middle. As a rough rule of thumb it is found that the 3 dB
bandwidth of the filter is about 1.5 times the frequency difference between the two
resonant frequencies. It is also found that for optimum performance the matching of the
filter is very important. To achieve this, matching resistors are often placed on the input
and output. If the filter is not properly matched then it is found that there will be more inband
ripple and the ultimate rejection may not be as good.

Diagram of half lattice crystal filter
A two pole filter (i.e. one with two crystals) is not normally adequate to meet many
requirements. The shape factor can be greatly improved by adding further sections.
Typically ultimate rejections of 70 dB and more are required in a receiver. As a rough
guide a two pole filter will generally give a rejection of around 20 dB; a four pole filter,
50 dB; a six pole filter, 70 dB; and an eight pole one 90 dB.
Crystal ladder filter
For many years the half lattice filter was possibly the most popular format used for
crystal filters. More recently the ladder topology has gained considerable acceptance. In
this form of crystal pass band filter all the resonators have the same frequency, and the
inter-resonator coupling is provided by the capacitors placed between the resonators with
the other termination connected to earth.
Four pole ladder crystal filter
Although crystal filters are widely used as the high performance filters in receivers,
mechanical filters are another option. Mechanical filters have been used for many years
and are able to provide excellent service at frequencies up to just under 1 MHz. These
filters can offer advantages over crystal filters in some instances being small, very stable,
and rugged. In fact they are not subject to deterioration due to continuous exposure to
shock and vibration, a factor that can be particularly important in some applications. A
further advantage is that they offer low levels of intermodulation distortion, a factor that
is often overlooked in many receiver designs.
Principles of operation
In essence their operation is very similar to that of a crystal, although the various
functions are performed by individual components within the filter. At either end of the
filter assembly there are transducers which convert the signals from their electrical form
to mechanical vibrations, and back again at the other end. These vibrations are applied to
a series of mechanical resonators which are mechanically resonant at the required
frequency. The resonators are mechanically coupled, typically with coupling wires to
transfer the vibrations from one section to the next. By altering the amount of coupling
between the sections and the natural frequency of each resonator, the response of the
overall unit can be tailored to meet the exact requirements.

Types of mechanical filter
There are several types of mechanical filter. The choice of the type depends upon the
frequency in use and the application.
The first type is known as the torsional filter. This type of mechanical filter uses rods that
vibrate in torsion. Electrical energy is coupled in by means of a piezoelectric ceramic
transducer into torsional motion. These filters are used for frequencies in the range from
below 100 kHz to just under 1 MHz.
Seven-resonator torsional mechanical filter
Image Courtesy Rockwell Collins
A second type of filter is known as the bar flexural mode mechanical filter. This is used
for low frequency designs, typically having a centre frequency between 5 to 100 kHz and
with bandwidths of .2 to 1.5 percent.
Bar flexural mode mechanical filter
Image Courtesy Rockwell Collins
Summary
Although mechanical filters are not used as widely as crystal filters, they can nevertheless
offer an excellent solution in some instances. In view of this they are the ideal solution
for many applications where high performance filters are required at frequencies below
about 700 kHz to 1MHz.
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SAW filters
- an overview or tutorial about SAW filters used as RF and IF filters.
Surface Acoustic Wave (SAW) technology is used in many areas of electronics to
provide resonators for oscillators, filters and transformers. One of the major uses of these
devices is as SAW filters which find widespread use in radio applications. These SAW

filters provide good performance filtering solutions while offering a cost effective
solution.
SAW filters are widely used in cell phone applications for filtering. Here they provide
considerable advantages in terms of cost and size, in an environment where these two
aspects are of considerable importance. Additionally their importance in the cellular
industry has meant that considerable amounts of research and development have been
undertaken on SAW filter s in the last 20 years, and their performance has improved
considerably in this period.
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DSP - Digital Signal Processing tutorial
- an overview or introduction to the basics of Digital Signal Processing, DSP, and
how it can be used in radio receiver technology to improve performance and
flexibility
Today, Digital Signal Processing, DSP, is widely used in radio receivers as well as in
many other applications from television, radio transmission, or in fact any applications
where signals need to be processed. Today it is not only possible to purchase digital
signal processor integrated circuits, but also DSP cards for use in computers. Using these
DSP cards it is possible to develop software or just use a PC platform in which to run the
DSP card.
DSP has many advantages over analogue processing. It is able to provide far better levels
of signal processing than is possible with analogue hardware alone. It is able to perform
mathematical operations that enable many of the spurious effects of the analogue
components to be overcome. In addition to this, it is possible to easily update a digital
signal processor by downloading new software. Once a basic DSP card has been
developed, it is possible to use this hardware design to operate in several different
environments, performing different functions, purely by downloading different software.
It is also able to provide functions that would not be possible using analogue techniques.
For example a complicated signal such as Orthogonal Frequency Division Multiplex
(OFDM) which is being used for many transmissions today needs DSP to become viable.
Despite this DSP has limitations. It is not able to provide perfect filtering, demodulation
and other functions. There are mathematical limitations. In addition to this the processing
power of the DSP card may impose some processing limitations. It is also more
expensive than many analogue solutions, and thus it may not be cost effective in some
applications. Nevertheless it has many advantages to offer, and with the wide availability
of cheap DSP hardware and cards, it often provides an attractive solution for many radio
applications.
What is DSP?
As the name suggests, digital signal processing is the processing of signals in a digital
form. DSP is based upon the fact that it is possible to build up a representation of the
signal in a digital form. This is done by sampling the voltage level at regular time
intervals and converting the voltage level at that instant into a digital number proportional
to the voltage. This process is performed by a circuit called an analogue to digital
converter, A to D converter or ADC. In order that the ADC is presented with a steady
voltage whilst it is taking its sample, a sample and hold circuit is used to sample the
voltage just prior to the conversion. Once complete the sample and hold circuit is ready to
update the voltage again ready for the next conversion. In this way a succession of
samples is made.

Sampling a waveform for DSP
Once in a digital format the real DSP is able to be undertaken. The digital signal
processor performs complicated mathematical routines upon the representation of the
signal. However to use the signal it then usually needs to be converted back into an
analogue form where it can be amplified and passed into a loudspeaker or headphones.
The circuit that performs this function is not surprisingly called a digital to analogue
converter, D to A converter or DAC.
Block diagram of a Digital Signal Processor, DSP)
The advantage of DSP, digital signal processing is that once the signals are converted
into a digital format they can be manipulated mathematically. This gives the advantage
that all the signals can be treated far more exactly, and this enables better filtering,
demodulation and general manipulation of the signal. Unfortunately it does not mean that
filters can be made with infinitely steep sides because there are mathematical limitations
to what can be accomplished.
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FPGAs for DSP Hardware
- the advantages and disadvantages of using FPGAs rather than DSP
processors in the DSP hardware.
When designing the hardware system for a DSP application it is necessary to carefully
consider the approach that will be taken. One of the fundamental decisions involves
whether to use a standard DSP processor, or whether to use an FPGA in the DSP
hardware. Each has its own advantages and they need to be carefully balanced at the
earliest stages of the design.
DSP processor
A DSP processor is a specialised processor that is designed specifically for operating

complex mathematically orientated intensive calculations very swiftly. As processing
needs to be undertaken almost in real time, the speed of the processor is one of the main
limiting performance criteria for the performance of the system For example very steep
filters need more processing than those that are not so steep, etc..
While DSP processors, despite their sophistication in terms of processing have
limitations, they also have advantages. One of these is in their cost. They may still be
expensive by some standards, but they are nevertheless cheaper than their counterparts,
the FPGA.
FPGAs for DSP
The other approach that many adopt is to use an FPGA as the core of the DSP hardware.
These devices can be programmed and there are many set cores that can be used to
provide the routines that are required. For example if a filter is required, then it is
possible to tailor circuitry within the FPGA to undertake this. Similarly other functions
can be programmed in on top of the basic processor. In this way the FPGA is able to be
programmed to provide a highly efficient and tailored solution.
The main disadvantage of the FPGA is its cost. FPGAs are more costly that DSP
processors and therefore performance has to be weighed against cost.
Summary
FPGAs and DSP processors provide two very different approaches to the design of DSP
hardware systems. Each have their own advantages. There are many high sampling rate
applications that an FPGA does easily, while the DSP could not. Equally, there are many
complex software problems that the FPGA cannot address.