Huang-Jen Chiu

Huang-Jen Chiu
Dept. of Electronic Engineering
National Taiwan University of
Science and Technology
Office: EE502-1
Tel: 02-2737-6419
E-mail: hjchiu@mail.ntust.edu.tw

Power Electronics
--Converters, Applications, and Design
Third Edition
Textbook
Mohan / Undeland / Robbins
民全書局 02-23657999 02-3651662
Midterm: 50% Final: 50%

� Power Electronic Systems
Outlines
� Overview of Power Semiconductor Switches
� Switch-Mode DC/DC Converters
� Switch-Mode DC/AC Inverters
� Resonant Converters
� Switching DC Power Supplies
� Power Conditioners and Uninterruptible Power Supplies
� Practical Converter Design Considerations

Chapter 1
Power Electronic Systems

Power Electronic Systems

Linear Power Supply
� Series transistor as an adjustable resistor
� Low Efficiency
� Heavy and bulky

Switch-Mode Switch Mode Power Supply
• Transistor as a switch
• High Efficiency
• High-Frequency Transformer

Basic Principle of
Switch-Mode Switch Mode Synthesis
• Constant switching frequency
• Pulse width controls the average
• L-C filters the ripple

Application
in Adjustable Speed Drives
• Conventional drive wastes energy across the
throttling valve to adjust flow rate
• Using power electronics, motor-pump speed is
adjusted efficiently to deliver the required flow rate

Scope and Applications

Scope and Applications

Classification of Power Converters
� ac-dc converters (controlled rectifiers)
� dc-dc converters (dc choppers)
� dc-ac converters (inverters)
� ac-ac converters (ac voltage controllers)

Power Processor as a
Combination of Converters
• Most practical topologies require an energy
storage element, which also decouples the input
and the output side converters

Power Flow through Converters
• Converter is a general term
• An ac/dc converter is shown here
• Rectifier Mode of operation when power from ac to dc
• Inverter Mode of operation when power from ac to dc

AC Motor Drive
• Converter 1 rectifies line-frequency ac into dc
• Capacitor acts as a filter; stores energy; decouples
• Converter 2 synthesizes low-frequency ac to motor
• Polarity of dc-bus voltage remains unchanged
– ideally suited for transistors of converter 2

Matrix Converter
• Very general structure
• Would benefit from bi-directional and bi-polarity switches
• Being considered for use in specific applications

Interdisciplinary Nature of
Power Electronics

Chapter 2 Overview of
Power Semiconductor Devices

Diodes
• On and off states controlled by the power circuit

Diode Turn-Off Turn Off
• Fast-recovery diodes have a small reverse-recovery time

Thyristors
• Semi-controlled device
• Latches ON by a gate-current pulse if forward biased
• Turns-off if current tries to reverse

Thyristor in a Simple Circuit
• For successful turn-off, reverse voltage required for
an interval greater than the turn-off interval

Generic Switch Symbol
• Idealized switch symbol
• When on, current can flow only in the direction of the arrow
• Instantaneous switching from one state to the other
• Zero voltage drop in on-state
• Infinite voltage and current handling capabilities

Switching Characteristics
(linearized)
Switching Power Loss is proportional to:
• switching frequency
1
V I
• turn-on and turn-off times s =
d o
P fs
(tc(on)
+ tc(off)
2
)

Bipolar Junction Transistors (BJT)
• Used commonly in the past
• Now used in specific applications
• Replaced by MOSFETs and IGBTs

Various Configurations of BJTs

MOSFETs
• Easy to control by the gate
• Optimal for low-voltage operation at high switching frequencies
• On-state resistance a concern at higher voltage ratings

Gate-Turn Gate Turn-Off Off Thyristors (GTO)
• Slow switching speeds
• Used at very high power levels
• Require elaborate gate control circuitry

GTO Turn-Off Turn Off
• Need a turn-off snubber

Insulated Gate Bipolar Transistor
(IGBT)

MOS-Controlled
MOS Controlled Thyristor
(MCT)
• Simpler Drive and faster switching speed than those of GTOs.
• Current ratings are significantly less than those of GTOs.

Comparison of Controllable Switches

Summary of Device Capabilities

Rating of Power Devices

Chapter 3
Review of Basic Electrical and
Magnetic Circuit Concepts

Sinusoidal Steady State
P
PF =
=
S
cosφ

Three-Phase Three Phase Circuit

Steady State in Power Electronics

Fourier Analysis
∞
1
f(t) = F0
+ ∑ fh
(t) = a0
+ ∑ h + h ω
2
∞
h=
1
h=
1
{ a cos(hωt)
b sin(h t) }

PF
Distortion in the Input Current
P I
= = s1 cosφ1
=
S I
s
DPF
• Voltage is assumed to be sinusoidal
1 + THD
• Subscript “1” refers to the fundamental
I
I
s1
s
=
DPF
• The angle is between the voltage and the current fundamental
1
2
i

Phasor Representation

Response of L and C
v
L =
L
di
dt
L
i
c =
C
dv
dt
c

Inductor Voltage and Current
in Steady State
• Volt-seconds over T equal zero.

Capacitor Voltage and Current
in Steady State
• Amp-seconds over T equal zero.

Ampere’s Ampere s Law
∫
H
dl
=
∑
• Direction of magnetic field due to currents
• Ampere’s Law: Magnetic field along a path
i

Direction of Magnetic Field
B =
μH

B-H H Relationship; Saturation
• Definition of permeability

Continuity of Flux Lines
φ + φ + φ =
0
1 2 3

Concept of Magnetic Reluctance
• Flux is related to ampere-turns by reluctance

Analogy between Electrical and
Magnetic Variables

Analogy between Equations in
Electrical and Magnetic Circuits

Faraday’s Faraday s Law and Lenz’s Lenz s Law
dφ
e = N =
dt
L
di
dt

Inductance L
• Inductance relates flux-linkage to current

Analysis of a Transformer

Transformer Equivalent Circuit

Including the Core Losses
L ' = (
l2
N
N
N
R 2'
=
(
N
1
2
1
2
)
)
2
2
R
L
2
l2

Chapter 4
Computer Simulation

System to be Simulated
• Challenges in modeling power electronic systems

Large-Signal Large Signal System Simulation
• Simplest component models

Small-Signal
Small Signal Linearized Model
for Controller Design
• System linearized around the steady-state point

Closed-Loop Closed Loop Operation:
Large Disturbances
• Simplest component models
• Nonlinearities, Limits, etc. are included

Modeling of Switching Operation
• Detailed device models
• Just a few switching cycles are studied

Modeling of a Simple Converter
⎡
⎢
⎢
⎢
⎣
di
dv dt
dt
L
c
⎤
⎥
⎥
⎥
⎦
=
r
i
L
L
⎡ rL
⎢
-
L
⎢ 1
⎢
⎣ C
di
iL
+ L
dt
dv
- C c -
dt
L
1
-
L
1
-
CR
+
vc
R
⎤
⎥
⎥
⎥
⎦
⎡
⎢
⎣
v
=
i
v
c
L
c
0
=
⎤
⎥
⎦
v
+
oi
⎡
⎢
⎢
⎣
0 L
1
⎤
⎥ v
⎥
⎦
oi

Modeling using PSpice
• Schematic approach is far superior

PSpice-based PSpice based Simulation
• Simulation results

Simulation using MATLAB

Chapter 5
Diode Rectifiers

Diode Rectifier Block Diagram
• Uncontrolled utility interface (ac to dc)

A Simple Circuit
• Resistive load

A Simple Circuit (R-L (R L Load)
• Current continues to flows for a while even
after the input voltage has gone negative

A Simple Circuit
(Load has a dc back-emf) back emf)
• Current begins to flow when the input voltage exceeds the dc back-emf
• Current continues to flows for a while even after the input voltage has
gone below the dc back-emf

Single-Phase Single Phase Diode Rectifier Bridge
• Large capacitor at the dc output for filtering and energy storage

Diode-Rectifier Diode Rectifier Bridge Analysis

Diode-Rectifier Diode Rectifier Bridge Input Current

Current Commutation
• Assuming inductance in this circuit to be zero

Current Commutation

Current Commutation
in Full-Bridge Full Bridge Rectifier

Current Commutation

Rectifier with a dc-side dc side voltage

Diode-Rectifier Diode Rectifier with a Capacitor Filter
• Power electronics load is represented by
an equivalent load resistance

Diode Rectifier Bridge
• Equivalent circuit for analysis on one-half cycle basis

Diode-Bridge Diode Bridge Rectifier: Waveforms
• Analysis using PSpice

Input Line-Current Line Current Distortion
• Analysis using PSpice

Line-Voltage Line Voltage Distortion
• PCC is the point of common coupling

Line-Voltage Line Voltage Distortion
• Distortion in voltage supplied to other loads

Voltage Doubler Rectifier
• In 115-V position, one capacitor at-a-time is
charged from the input.

A Three-Phase, Three Phase, Four-Wire Four Wire System
• A common neutral wire is assumed

Three-Phase, Three Phase, Full-Bridge Full Bridge Rectifier
• Commonly used

Three-Phase, Three Phase, Full-Bridge Full Bridge Rectifier
• Output current is assumed to be dc

Three-Phase, Three Phase, Full-Bridge Full Bridge Rectifier:
Input Line-Current
Line Current
• Assuming output current to be purely dc and
zero ac-side inductance

Rectifier with a Large Filter Capacitor
• Output voltage is assumed to be purely dc

Chapter 6
Thyristor Converters
• Controlled conversion of ac into dc

Chapter 6
Thyristor Converters
• Controlled conversion of ac into dc

Thyristor Converters
• Two-quadrant conversion

Primitive circuits with thyristors

Thyristor Triggering

Full-Bridge Full Bridge Thyristor Converters
• Single-phase and three-phase

Single-Phase Single Phase Thyristor Converters

Average DC Output Voltage
is s1
s3 s1
( ωt)
= 2I
sin( ωt
- ∂ ) + 2I
I sin[3( ωt
- ∂ )]
2
I s1 = 2I
d =
π
0.9I
• Assuming zero ac-side inductance
d
⇒ P
= 0.9cos
+
∂
...

Input Line-Current Line Current Waveforms
• Harmonics, power and reactive power

1-Phase Phase Thyristor Converter

Thyristor Converter

DC Voltage versus Load Current
• Various values of delay angle

Thyristor Converters:
Inverter Mode
• Assuming the ac-side inductance to be zero

Thyristor Converters:
Inverter Mode
• Family of curves at various values of delay angle

Thyristor Converters:
Inverter Mode

Thyristor Converters:
Inverter Mode

3-Phase Phase Thyristor Converters

Chapter 7
DC-DC DC DC Switch-Mode Switch Mode Converters
• dc-dc converters for switch-mode dc power supplies and
dc-motor drives

Block Diagram of DC-DC DC DC Converters
• Functional block diagram

Stepping Down a DC Voltage
• A simple approach that shows the evolution

Pulse-Width Pulse Width Modulation in
DC-DC DC DC Converters

d
o
Step-Down Step Down DC-DC DC DC Converter
( V − V ) T = V
Vo on
V
d
on
T
= = D
T
o
< 1
T
off

Waveforms at the boundary of
Cont./ Discont. Discont.
Conduction
1 t
T V
I I on (V -V
) s d
LB =
L, peak = d o = D(1-
D) = 4ILB,
maxD(1-
D)
2 2L
2L
• Critical current below which inductor current becomes
discontinuous

Step-Down Step Down DC-DC DC DC Converter:
Discontinuous Conduction Mode
• Steady state; inductor current discontinuous
V
V
o
d
=
D
2
+
1
4
D
(
I
2
I
o
LB, max
)

Limits of Cont./ Discont. Discont.
Conduction
V
V
o
d
=
D
2
+
1
4
D
(
I
2
V
V
I
o =
d
o
LB, max
D : CCM
)
:
DCM

ΔQ
Δ Vo
=
=
C
Output Voltage Ripple
ΔI
LT
8C
s

d
Step-Up Step Up DC-DC DC DC Converter
V T = ( V −V
) T
= > 1
on
o
d
off
• Output voltage must be greater than the input
V
V
o
d
1
1
− D

Limits of Cont./ Discont. Discont.
Conduction
1 t T V
I I on V s o
LB = L, peak = d = D(1-
D) = 4ILB,
maxD(1-
D)
2 2L 2L
TsVo
2 27 2
I oB =
(1-
D)ILB
= D(1-
D) = D(1-
D) IoB,
max
2L
4

D =
4
27
V
V
o
d
V
(
V
Discont. Discont.
Conduction
o
d
-1)
I
I
o
oB, max

V
V
o
d
Limits of Cont./ Discont. Discont.
1
= : CCM
1−
D
Conduction
D =
4
27
V
V
o
d
V
(
V
o
d
-1)
I
I
o
oB, max
: DCM

Output Ripple
ΔV
o
=
I
o
t
C
on
=
V
o
R
DT
C
s

Step-Down/Up Step Down/Up DC-DC DC DC Converter
V T =
d
on
V
o
T
off
D
− D
• The output voltage can be higher or lower than
the input voltage
V
V
o
d
= 1

Limits of Cont./ Discont. Discont.
Conduction
1 t T V
I I on V s o
LB = L, peak = d = (1-
D) = ILB,
max(1-
D)
2 2L 2L
TsVo
2
2
I oB =
(1-
D)ILB
= (1-
D) = IoB,
max(1-
D)
2L

Discontinuous Conduction Mode
V
D =
V
o
d
I
I
o
oB, max
• This occurs at light loads

V o
V d
Limits of Cont./ Discont. Discont.
= 1
D
− D
:
Conduction
CCM
V
D =
V
o
d
I
I
o
oB, max
: DCM

Output Voltage Ripple
• ESR is assumed to be zero
ΔV
o
=
I
o
t
C
on
=
V
o
R
DT
C
s

• The output voltage can
be higher or lower than
the input voltage
Cuk DC-DC DC DC Converter

Converter for DC-Motor DC Motor Drives

Converter Waveforms

Output Ripple in Converters for
DC-Motor DC Motor Drives

Switch Utilization
in DC-DC DC DC Converters
• It varies significantly in various converters

Reversing the Power Flow
in DC-DC DC DC Converters

Chapter 8
Switch-Mode Switch Mode DC-AC DC AC Inverters
• Converters for ac motor drives and
uninterruptible power supplies

Switch-Mode Switch Mode DC-AC DC AC Inverter

Switch-Mode Switch Mode DC-AC DC AC Inverter

V
m =
a
m =
f
Synthesis of a Sinusoidal Output
^
control
^
Vtri
f
f
s
1
by PWM

Details of a Switching Time Period
• Small m f (m f ≤21): Synchronous PWM
• Large m f (m f >21): Asynchronous PWM

Harmonics in the DC-AC DC AC Inverter
Output Voltage
• Harmonics appear around the carrier frequency and its multiples

Harmonics due to Over-modulation
Over modulation
• These are harmonics of the fundamental frequency

Square-Wave Square Wave Mode of Operation
• Harmonics are of the fundamental frequency
• Less switching losses in high power applications
• The DC input voltage must be adjusted

Half-Bridge Half Bridge Inverter
• Capacitors provide the mid-point

Single-Phase Single Phase Full-Bridge Full Bridge DC-AC DC AC Inverter
• Consists of two inverter legs

PWM to Synthesize Sinusoidal Output

Analysis assuming Fictitious Filters
• Small fictitious filters eliminate the switching-frequency
related ripple

DC-Side DC Side Current

Uni-polar Uni polar Voltage Switching

DC-Side DC Side Current
in a Single-Phase Single Phase Inverter

Sinusoidal Synthesis by Voltage Shift
• Phase shift allows voltage cancellation to synthesize a
1-Phase sinusoidal output

Square-Wave Square Wave and PWM Operation
• PWM results in much smaller ripple current

Push-Pull Push Pull Inverter
• Only one switch conducts at any instant of time
• High efficiency for low-voltage source applications

Three-Phase Three Phase Inverter
• Three inverter legs; capacitor mid-point is fictitious

Three-Phase Three Phase PWM Waveforms

Three-Phase Three Phase Inverter Harmonics

Three-Phase Three Phase Inverter Output

Square-Wave Square Wave and PWM Operation
• PWM results in much smaller ripple current

DC-Side DC Side Current
in a Three-Phase Three Phase Inverter
• The current consists of a dc component and the
switching-frequency related harmonics

Effect of Blanking Time
• Results in nonlinearity

Effect of Blanking Time
ΔV
o
⎧ 2t
⎪ T
= s
⎨
2t
⎪-
⎪⎩
T
> 0
< 0
• Voltage jump when the current reverses direction
Δ
Δ
s
V
d
V
d
, i
o
, i
o

Effect of Blanking Time
• Effect on the output voltage

Programmed Harmonic Elimination
• Angles based on the desired output

Tolerance-Band Tolerance Band Current Control
• Results in a variable frequency operation

Fixed-Frequency Fixed Frequency Operation
• Better control is possible using dq analysis

Chapter 9
Zero-Voltage Zero Voltage or Zero-Current
Zero Current Switchings
• converters for soft switching

Hard Switching Waveforms
• The output current can be positive or negative

Turn-on Turn on and Turn-off Turn off Snubbers

Switching Trajectories
• Comparison of Hard versus soft switching

Undamped
Undamped Series
Series-Resonant Circuit
Resonant Circuit
L
c
r
d
c
L
r
i
dt
dv
C
V
v
dt
di
L
=
=
+
)
t
t
(
sin
I
Z
)
t
-
(t
)cos
V
-
(V
-
V
(t)
v
)
t
t
(
sin
Z
V
-
V
)
t
-
(t
cos
I
(t)
i
o
o
Lo
o
o
o
co
d
d
c
o
o
o
co
d
o
o
Lo
L
−
+
=
−
+
=
ω
ω
ω
ω
V d

Series
Series-Resonant Circuit
Resonant Circuit
with Capacitor
with Capacitor-Parallel Load
Parallel Load
o
L
c
r
c
d
c
L
r
I
-
i
dt
dv
C
i
V
v
dt
di
L
=
=
=
+
)
t
t
(
sin
)
I
-
(I
Z
)
t
-
(t
)cos
V
-
(V
-
V
(t)
v
)
t
t
(
sin
Z
V
-
V
)
t
-
(t
)cos
I
-
(I
I
(t)
i
o
o
o
Lo
o
o
o
co
d
d
c
o
o
o
co
d
o
o
o
Lo
o
L
−
+
=
−
+
+
=
ω
ω
ω
ω

Impedance of a Series-Resonant Series Resonant Circuit
Q
=
ω
L
R
1
C
o r = =
ω o r R
Z
R
o
• The impedance is capacitive below the resonance frequency

Undamped
Undamped Parallel
Parallel-Resonant Circuit
Resonant Circuit
dt
di
L
v
I
dt
dv
C
i
L
r
c
d
c
r
L
=
=
+
)
t
t
(
cos
V
)
t
-
(t
)sin
I
-
(I
Z
(t)
v
)
t
t
(
sin
Z
V
)
t
-
(t
)cos
I
-
(I
I
(t)
i
o
o
o
c
o
o
Lo
d
o
c
o
o
o
co
o
o
d
Lo
d
L
−
+
=
−
+
+
=
ω
ω
ω
ω

Impedance of a Parallel-Resonant Parallel Resonant Circuit
R
Q = ω
o RC r = =
ω L
o
r
R
Z
o
• The impedance is inductive at below the resonant frequency

Series-Loaded Series Loaded Resonant (SLR) Converter
2ωs

SLR Converter Waveforms
1/2ωo

ZVS, ZCS
SLR Converter Waveforms
ωs >ωo Turn
Large
on
with
turn - off
ZVS
and
switching
Controllable
switches used
ZCS
losses

Lossless Snubbers in SLR Converters
• The operating frequency is above the resonance frequency

SLR Converter Characteristics
• The operating frequency is varied to regulate the output voltage

SLR Converter Control
• The operating frequency is varied to regulate the output voltage

ZCS
Parallel-Loaded Parallel Loaded Resonant (PLR) Converter
No turn - on and turn - off
losses
ZVS, ZCS
ω ≤
s
1
ω
2
o

No
PLR Converter Waveforms
turn - off
losses
ZVS, ZCS
1
ω o < ω s <
2
ω
o

No
ZVS
PLR Converter Waveforms
turn
- on
losses

PLR Converter Characteristics
• Output voltage as a function of operating frequency
for various values of the output current

Hybrid-Resonant Hybrid Resonant DC-DC DC DC Converter
• Combination of series- and parallel-loaded resonances
• A SLR offers an inherent current limiting under short-circuit conditions and
a PLR regulating its voltage at no load with a high-Q resonant tank is not a
problem

Parallel-Resonant
Parallel Resonant
Current-Source Current Source Converter
Induction
Coil
Resistive
Capacitive
• Basic circuit to illustrate the operating principle at the
fundamental frequency

Parallel-Resonant
Parallel Resonant
Current-Source Current Source Converter
• Using thyristors; for induction heating

Single-switch
ZCS Turn-on
Class-E Class E Converters
ZVS Turn-off
Used
for
electronic
Sin-wave Current
High
No
peak
high
-
ballasts
switching
volatge
frequency
and
losses
current

Class-E Class E Converters

Resonant Switch Converters

ZCS Turn-on
ZCS Resonant-Switch Resonant Switch Converter
ZCS Turn-off
Voltage is regulated by varying
the switching frequency

ZCS Turn-on
ZCS Resonant-Switch Resonant Switch Converter
Accelerating diode
ZCS Turn-off
Discharge slowly at light load

ZVS Resonant-Switch Resonant Switch Converter
ZVS Turn-off
ZVS Turn-on

MOSFET Internal Capacitances
ZVS is preferable over ZCS at
high switching frequencies
• These capacitances affect the MOSFET switching

ZVS-CV ZVS CV DC-DC DC DC Converter
ZVS Turn-on
• The inductor current must reverse direction
during each switching cycle

ZVS-CV ZVS CV DC-DC DC DC Converter

ZVS-CV ZVS CV Principle Applied to
DC-AC DC AC Inverters

Three-Phase Three Phase ZVS-CV ZVS CV DC-AC DC AC Inverter
• Very large ripple in the output current

Output Regulation by Voltage Control
• Each pole operates at nearly 50% duty-ratio

ZVS-CV ZVS CV with Voltage Cancellation
• Commonly used

Resonant DC-Link DC Link Inverter
• The dc-link voltage is made to oscillate
ZVS Turn-on

Three-Phase Three Phase Resonant DC-Link DC Link Inverter
• Modifications have been proposed

High-Frequency
High Frequency-Link Link Inverter
• Basic principle for selecting integral half-cycles of
the high-frequency ac input

High-Frequency
High Frequency-Link Link Inverter
• Low-frequency ac output is synthesized by selecting
integral half-cycles of the high-frequency ac input

High-Frequency
High Frequency-Link Link Inverter
• Shows how to implement such an inverter

Chapter 10
Switching DC Power Supplies
• One of the most important applications of power electronics

Linear Power Supplies
• Very poor efficiency and large weight and size

Switching DC Power Supply
• High efficiency and small weight and size

Switching DC Power Supply:
Multiple Outputs
• In most applications, several dc voltages are required,
possibly electrically isolated from each other

Transformer Analysis
• Needed to discuss high-frequency isolated supplies

PWM to Regulate Output

Flyback Converter
• Derived from buck-boost; very power at small power
(> 50 W ) power levels

Flyback Converter
• Switch on and off states (assuming incomplete
core demagnetization)

Flyback Converter
• Switching waveforms (assuming incomplete
core demagnetization)

Other Flyback Converter Topologies

Forward Converter
• Derived from Buck; idealized to assume that the
transformer is ideal (not possible in practice)

Forward Converter: in Practice
• Switching waveforms (assuming incomplete
core demagnetization)

Forward Converter:
Other Possible Topologies
• Two-switch Forward converter is very commonly used

Push-Pull Push Pull Inverter
• Leakage inductances become a problem

Half-Bridge Half Bridge Converter
• Derived from Buck

Full-Bridge Full Bridge Converter
• Used at higher power levels (> 0.5 kW )

Current-Source Current Source Converter
• More rugged (no shoot-through) but both switches must
not be open simultaneously

Ferrite Core Material
• Several materials to choose from based on applications

Core Utilization in Various
Converter Topologies
• At high switching frequencies, core losses limit excursion
of flux density

Control to Regulate Voltage Output
• Linearized representation of the feedback control system

⎪⎩
⎪
⎨
⎧
−
+
=
+
=
•
•
s
d
s
d
T
d
v
B
x
A
x
dT
v
B
x
A
x
)
1
(
,
,
2
2
1
1
⎩
⎨
⎧
−
=
=
s
o
s
o
T
d
x
C
v
dT
x
C
v
)
1
(
,
,
2
1
⎪⎩
⎪
⎨
⎧
−
+
=
−
+
+
−
+
=
⇒
•
x
d
C
d
C
v
v
d
B
d
B
x
d
A
d
A
x
o
d
)]
1
(
[
)]
1
(
[
)]
1
(
[
2
1
2
1
2
1
d
V
d
D
B
d
D
B
x
X
d
D
A
d
D
A
x
X )]
(
1
[
)
(
[
)
)]}(
(
1
[
)
(
{
~
2
~
1
~
~
2
~
1
~
+
−
+
+
+
+
+
−
+
+
=
+
•
•
d
V
d
B
D
B
d
B
D
B
x
X
d
A
D
A
d
A
D
A ]
)
1
(
[
)
](
)
1
(
[
~
2
2
~
1
1
~
~
2
2
~
1
1
−
−
+
+
+
+
−
−
+
+
=
~
~
2
1
~
2
1
~
2
1
2
1
2
1
2
1
)
(
)]
1
(
[
]
)
(
)
[(
)]
1
(
[
)]
1
(
[
x
d
A
A
x
D
A
D
A
d
V
B
B
X
A
A
V
D
B
D
B
X
D
A
D
A d
d
−
+
−
+
+
−
+
−
+
−
+
+
−
+
=
Linearization of the Power Stage
Linearization of the Power Stage

Linearization of the Power Stage
Linearization of the Power Stage
~
2
1
2
1
~
~
]
)
(
)
[( d
V
B
B
X
A
A
x
A
BV
AX
x
X d
d
−
+
−
+
+
+
≈
+
•
•
~
2
1
2
1
~
~
]
)
(
)
[( d
V
B
B
X
A
A
x
A
x d
−
+
−
+
=
⇒
•
d
BV
AX
X +
=
=
•
0
Θ
~
~
2
1
~
2
1
~
2
1
2
1
~
~
2
~
1
~
)
(
)]
1
(
[
]
)
[(
)]
1
(
[
]
)][
(
1
[
)
(
{
d
x
C
C
x
D
C
D
C
d
X
C
C
X
D
C
D
C
x
X
d
D
C
d
D
C
v
V o
o
−
+
−
+
+
−
+
−
+
=
+
+
−
+
+
=
+
~
~
2
1
~
]
)
[( x
C
d
X
C
C
CX
v
V o
o
+
−
+
≈
+
CX
V o =
Θ
~
2
1
~
~
]
)
[( d
X
C
C
x
C
v o
−
+
=
⇒

•
Linearization of the Power Stage
X = 0 = AX + BV
and Vo
= CX
~
~
d
V
⇒
V
o
d
= −CA
−1
B
•
~ ~
x = Ax+
A1
− A2
) X + ( B1
−B2
) Vd
Steady-state
[( ] d
⇒s
x(
s)
= Ax(
s)
+ [( A1
− A2
) X + ( B1
−B2
) Vd
] d(
s)
~
−1
⇒ x( s)
= [ sI − A]
[( A1
− A2
) X + ( B1
−B2
) Vd
] d(
s)
⇒
~
DC voltage transfer ratio
vo(
s)
−1
( s)
= = C[
sI − A]
[( A1
− A2
) X + ( B1
−B2
) V ] + ( C1
−C2)
X
~
d(
s)
Tp d
~
~
~
~
vo ~
=
Cx+
[( C −C
) X]
d
1
2
~

Forward Converter: An Example
Forward Converter: An Example
⎪⎩
⎪
⎨
⎧
=
−
+
+
−
=
−
+
+
+
−
•
•
•
•
0
)
(
0
)
(
2
1
2
2
2
1
1
1
x
C
x
R
x
Cr
x
x
C
x
R
x
r
x
L
V
c
L
d
d
c
c
c
c
L
c
L
c
V
L
x
x
r
R
C
r
R
C
R
r
R
L
R
r
R
L
r
r
Rr
Rr
x
x
⎥
⎥
⎦
⎤
⎢
⎢
⎣
⎡
+
⎥
⎦
⎤
⎢
⎣
⎡
⎥
⎥
⎥
⎥
⎦
⎤
⎢
⎢
⎢
⎢
⎣
⎡
+
−
+
+
−
+
+
+
−
=
⎥
⎥
⎦
⎤
⎢
⎢
⎣
⎡
•
•
0
1
)
(
1
)
(
)
(
)
(
2
1
2
1
A 1 =A 2 B1
B 2 =0

R >> r + ) ⇒
( C rL
v
o
= R(
x
1
1
•
−C
x
2
⎡ Rrc
) = ⎢
⎣R
+ r
1
c
R
R+
r
⇒ A = A , B = B D , C = C
A=
A
1
=
A
2
⎡ rc
+ r
⎢−
≈ L
⎢ 1
⎢
⎣ C
L
1 ⎤
−
L ⎥
1 ⎥
− ⎥
CR⎦
1
c
⎤⎡x1
⎤
⎥⎢
⎥
⎦⎣x2⎦
C 1 =C 2
C = C = C ≈
1 2 c
[ r 1]
⎡1/
L⎤
B = B1D
= ⎢ ⎥D ⎣ 0 ⎦

⎥
⎥
⎥
⎦
⎤
⎢
⎢
⎢
⎣
⎡
+
−
−
−
+
+
=
−
L
r
r
C
L
CR
R
r
r
LC
A
L
c
L
c
1
1
1
/
)
(
1
1 D
r
r
R
r
R
D
V
V
L
c
c
d
o ≈
+
+
+
=
⇒
)
(
{ } 2
2
2
2
2
1
2
1
2
1
1
~
~
2
/
1
]
/
)
(
/
1
[
1
)
(
]
)
(
)
[(
]
[
)
(
)
(
)
(
o
o
z
z
o
d
L
c
c
d
d
o
p
s
s
s
V
LC
L
r
r
CR
s
s
LC
C
sr
V
X
C
C
V
B
B
X
A
A
A
sI
C
s
d
s
v
s
T
ω
ξω
ω
ω
ω
+
+
+
=
+
+
+
+
+
≈
−
+
−
+
−
−
=
=
−

Forward Converter:
Transfer Function Plots
T
p
( s)
=
V
d
2
ωo
ω
z
s
2
s+
ωz
+ 2ξω
s+
ω
o
2
o

Flyback Converter:
Transfer Function Plots
T
p
( 1+
s/
ωz1
)( 1−s
/ ωz2)
( s)
= Vd
f ( D)
2
as + b s+
c
o

Linearizing the PWM Block
~
d(
s)
1
vo(
s)
vo(
s)
d(
s)
T m(
s)
= = ⇒T
( ) ( )
~ ^
l ( s)
= = = T s T s
~ ~ ~ p m
v ( s)
V
v ( s)
d(
s)
v ( s)
c
r
~
c
~
~
c

Typical Gain and Phase Plots of the
Open-Loop Open Loop Transfer Function
• Definitions of the crossover frequency, phase and gain margins

A General Amplifier for
Error Compensation
• Can be implemented using a single op-amp

Type-2 Type 2 Error Amplifier
• Shows phase boost at the crossover frequency

Feedback-Loop Feedback Loop Stabilization

Feedback-Loop Feedback Loop Stabilization
F
F
co K =
=
z
F
F
p
co

Feedback-Loop Feedback Loop Stabilization
θ
total
lag
= 270 ° − tan
K
+
tan
−1
−1
F
F
co K = =
1
K
z
F
F
p
co

L
C
F
F
o
o
o
esr
=
Compensator Design Example
3V
I
o
on
T
= 65×
10
=
2π
1
L
=
−6
o
esr
�� Vo 5V
�� Io(nom) o(nom)
10A
�� Io(min) o(min)
1A
�� Switching frequency 100kHz
�� Minimum output ripple 50mV
C
3 × 5 × 10 × 10
10
dI
V
o
or
o
=
= 65×
10
806
Hz
−6
−6
×
= 15 μH
2
0.
05
1
1
6
2 R C 2 65 10
=
= =
−
π
π × ×
= 2600μF
2.
5
kHz
50mV P-P

Compensator Design Example
G
m
=
G m + G s
0.
5(
V
sp
3
− 1)
=
0.
5 × ( 11 − 1)
3
= + 4.
5dB
2.
5
= 4. 5 + 20 log( ) = 4.
5 − 6 = −1.
5dB
5
R = R × 100(
40dB)
= 1k
× 100 = 100kΩ
2
1

F
1
Fco = Fs
5
z
F
F
co
esr
EA
=
Compensator Design Example
20 k
= = 8 ⇒ lag = 97°
2.
5k
F
K
lag
co
=
F p = K × Fco
=
20 kHz
= 360 − 45 − 97 = 218 ° ⇒ K =
20
4
=
= 5kHz
⇒
C1
=
4
1
2π
( 100 k )( 5k
)
=
318
pF
1
4 × 20 = 80 kHz ⇒ C 2 =
=
2π
( 100 k )( 80 k )
20
pF

Voltage Feed-Forward
Feed Forward
• Makes converter immune from input voltage variations

Voltage versus Current Mode Control

Various Types of Current Mode Control

Peak Current Mode Control
• Slope compensation is needed

A Typical PWM Control IC

Current Limiting

Implementing Electrical Isolation
in the Feedback Loop

Implementing Electrical Isolation
in the Feedback Loop

Input Filter
• Needed to comply with the EMI and harmonic limits

ESR of the Output Capacitor
• ESR often dictates the peak-peak voltage ripple

Chapter 11
Power Conditioners and
Uninterruptible Power Supplies
• Becoming more of a concern as utility de-regulation proceeds

Distortion in the Input Voltage
• The voltage supplied by the utility may not be sinusoidal

Typical Voltage Tolerance
Envelope for Computer Systems
• This has been superceded by a more recent standard

Typical Range of Input Power Quality

Electronic Tap Changers
• Controls voltage magnitude by connecting the output to
the appropriate transformer tap

Uninterruptible Power Supplies
(UPS)
• Block diagram; energy storage is shown to be in
batteries but other means are being investigated

UPS: Possible Rectifier Arrangements
• The input normally supplies power to the load as well
as charges the battery bank

UPS: Another Possible Rectifier
Arrangement
• Consists of a high-frequency isolation transformer

UPS: Another Possible Input
Arrangement
• A separate small battery charger circuit

Battery Charging Waveforms as
Function of Time
• Initially, a discharged battery is charged with a constant current

UPS: Various Inverter Arrangements
• Depends on applications, power ratings

UPS: Control
• Typically the load is highly nonlinear and the voltage output
of the UPS must be as close to the desired sinusoidal
reference as possible

UPS Supplying Several Loads
• With higher power UPS supplying several loads,
malfunction within one load should not disturb the other
loads

Another Possible UPS Arrangement
• Functions of battery charging and the inverter are combined

UPS: Using the Line Voltage as Backup
• Needs static transfer switches

Chapter 16
Residential and Industrial Applications
• Significant in energy conservation; productivity

Inductive Ballast of Fluorescent Lamps
• Inductor is needed to limit current

Rapid-Start Rapid Start Fluorescent Lamps
• Starting capacitor is needed

Electronic Ballast for Fluorescent Lamps
• Lamps operated at ~40 kHz

Induction Cooking
• Pan is heated directly by circulating currents – increases
efficiency

Industrial Induction Heating
• Needs sinusoidal current at the desired frequency: two options

Welding Application

Switch-Mode Switch Mode Welders
• Can be made much lighter weight

Chapter 17
Electric Utility Applications
• These applications are growing rapidly

HVDC Transmission
• There are many such systems all over the world

Control of HVDC Transmission System
• Inverter is operated at the minimum extinction angle
and the rectifier in the current-control mode

HVDC Transmission: AC-Side AC Side Filters
Tuned for the lowest (11 th and the 13 th harmonic)
frequencies

Effect of Reactive Power on
Voltage Magnitude

Thyristor-Controlled Thyristor Controlled Inductor (TCI)
• Increasing the delay angle reduces the reactive power
drawn by the TCI

Thyristor-Switched Thyristor Switched Capacitors (TSCs ( TSCs)
• Transient current at switching must be minimized

Instantaneous VAR Controller (SATCOM)
• Can be considered as a reactive current source

Characteristics of Solar Cells
• The maximum power point is at the knee of the characteristics

Photovoltaic Interface
• This scheme uses a thyristor inverter

Harnessing of Wing Energy
• A switch-mode inverter may be needed on
the wind generator side also

Active Filters for Harmonic Elimination
• Active filters inject a nullifying current so that the current
drawn from the utility is nearly sinusoidal

Chapter 18
Utility Interface
• Power quality has become an important issue

Various Loads Supplied by
the Utility Source
• PCC is the point of common coupling

Diode-Rectifier Diode Rectifier Bridge

Typical Harmonics in the Input Current
• Single-phase diode-rectifier bridge

Harmonic Guidelines: IEEE 519
• Commonly used for specifying limits on the input current
distortion

Harmonic Guidelines: IEEE 519
• Limits on distortion in the input voltage supplied by the utility

Reducing the Input Current Distortion
• use of passive filters

Power-Factor
Power Factor-Correction Correction (PFC) Circuit
• For meeting the harmonic guidelines

Power-Factor
Power Factor-Correction Correction (PFC)
Circuit Control
• generating the switch on/off signals

Power-Factor
Power Factor-Correction Correction (PFC) Circuit
• Operation during each half-cycle

Switch-Mode Switch Mode Converter Interface
• Bi-directional power flow; unity PF is possible

Switch-Mode Switch Mode Converter Control
• DC bus voltage is maintained at the reference value

Switch-Mode Switch Mode Converter Interface

EMI: Conducted Interefence
• Common and differential modes

Switching Waveforms
• Typical rise and fall times

Conducted EMI
• Various Standards

Conducted EMI Filter

V d
-
+
i D F
D f
Turn-off Turn off Snubber
I o
D s
Turn-off
snubber
I o - i
S R s
i
w i sw
C
C C s
s
s
V d
C s = I o t fi
2V d , ton>2.3RsCs, Vd/Rs

+
V d
-
D f
L s
S w
I o
R Ls
D Ls
Turn-on Turn on Snubber
D f
Snubber
circuit
+
V d
-
L s
D f
R Ls
D Ls
S w
Δv sw = L s I o
t ri toff>2.3Ls/Rs Pr=1/2LsIo^2fs
I o
I o
i
sw
With
snubber
Without
snubber
di
L sw
s
dt
V d
v sw

Aspects of EMC (EMI、EMS)
(EMI EMS)
�� EMC is concerned with the generation,
transmission, and reception of
electromagnetic energy
�� EMI occurs if the received energy
causes the receptor to behave in an
undesired manner

EMI Sources and Sensors

Three Ways to Prevent Interference
� Suppress the emission at its source
� Make the coupling path as inefficient as
possible
� Make the receptor less susceptible to
the emission

Four Basic EMC Problems

Other Aspects of EMC

EMC Requirements
�� Those required by governmental agencies
�� Those imposed by the product manufacturer

Frequency Range of
EMC Requirements

National Regulations Summary

Federal Communications
Commission (FCC)
�� Class A – for use in a commercial, industrial
or business environment
�� Class B – for use in a residential
environment

FCC Emission for Class B

FCC Emission for Class A

Comparison of the FCC Class A and
Class B Radiated Emission Limits

Open Area Test Site

Chamber for Measurement of
Radiated Emissions

Radiated EMI Test Setup

Antennas

Conducted EMI Test Setup

Line Impedance Stabilization Network
(LISN)

Conducted Emissions Test Layout

Conducted Emissions Test Layout

CISPR Bandwidth Requirements

Envelope
Detector
Quasi-Peak
Detector
Average
Detector
Three Detection Modes

Design Constraints for Products
� Product Cost
� Product Marketability
� Product Manufacturability
� Product Development Schedule

Advantages of EMC Design
� Minimizing the additional cost required
by suppression elements or redesign
� Maintaining the development and product
announcement schedule
� Insuring that the product will satisfy
the regulatory requirements

Effects of Component Leads

Resistors

1000Ω, Carbon Resistor
having 1/4 Inch Lead Lengths

Capacitors

470 pF Ceramic Capacitor with
Short Lead Lengths

470 pF Ceramic Capacitor with
1/2 Inch Lead Lengths

0.15 μF Tantalum Capacitor with
Short Lead Lengths

0.15 μF Tantalum Capacitor with
1/2 Inch Lead Lengths

Inductors

1.2μH Inductor

Common-Mode Common Mode Choke

Common-Mode Common Mode Choke

Frequency Response of the
Relative Permeabilities of Ferrite

Ferrite Beads

Multi-Turn Multi Turn Ferrite Beads

Driver Circuit of the DC Motor

The Periodic, Trapezoidal Pulse
Train Representing Clock and
Data Signals
The key parameters that contribute to the highfrequency
spectral content of the waveform are the
rise-time and fall-time of the pulse.

The Spectra of 1V, 10MHz,
50% Duty Cycle Trapezoidal Pulse Trains
for Rise-/Fall-time of 20ns/5ns

Spectrum Analyzer

The Effect of Bandwidth on Spectrum

The Effects of Differential-Mode
Current and Common-Mode Currents
� Common-mode current often produce larger radiated
emissions than the differential-mode currents

Differential-Mode Current Emission
E , max
| | =
I
D
Kf
D 2
A

Radiated Emission due to
the Differential-Mode Currents

Common Mistakes that Lead to
Unnecessarily Large DM Emissions

Common-Mode Current Emission
|
E
C
I
,
max
C
| =
Kf L

Radiated Emission due to
the Common-Mode Currents

Susceptibility Models

10V/m, 100MHz Incident
Uniform Plane Wave

Measurement of Conducted Emissions

Line Impedance Stabilization Network
(LISN)

Differential-Mode and Common-Mode
Current Components

Methods of Reducing the Common-Mode
Conducted Emissions

Definition of the Insertion Loss
of a Filter

Four Simple Filters
V L , wo
ω
L
IL = 20 log ( )
10 = 20 log ( )
10
V
R + R
L , w
S
L

Insertion Loss Tests

Conducted EMI Filter

Common-Mode Common Mode Choke

The Equivalent Circuit of the Filter
for Common-Mode Common Mode Currents

The Equivalent Circuit of the Filter
for Differential-Mode Differential Mode Currents

The Dominant Component of
Conducted Emission
^
I
^
^
Total =
I C ± I
D

A Device to Separate the CM
and DM Conducted Emissions

Measured Conducted Emissions
without Power Supply Filter

Measured Conducted Emissions
with 3300pF 3300pF
Line-to Line to-Ground Ground Cap.

Measured Conducted Emissions
with a 0.1μF 0.1 F Line-to Line to-Line Line Cap.

Measured Conducted Emissions
with a Green Wire Inductor

Measured Conducted Emissions
with a Common-Mode Common Mode Choke

Nonideal Effects in Diodes

Construction of Transformers

The Effect of Primary-to
Primary to-Secondary
Secondary
Capacitance of a Transformer

The Proper Filter Placement in the
Reduction of Conducted Emissions

Crosstalk
� The unintended EM coupling between wires and
PCB lands that are in close proximity.
� Crosstalk between wires in cables or between lands
on PCBs concerns the intrasystem interference
performance of the product.

Three-Conductor Three Conductor Transmission
Line illustrating Crosstalk

Wire-type Wire type Line illustrating Crosstalk

PCB Transmission Lines
illustrating Crosstalk

The Equivalent Circuit of TEM Wave
on Three-Conductor Three Conductor Transmission Line

The Simple Inductive-Capacitive
Inductive Capacitive
Coupling Model

Frequency Response of the Crosstalk
^
NE
^
V S
V
^
FE
^
V S
V
=
=
=
=
jω(
R
NE
Transfer Functions
R
+
NE
R
FE
IND
j ω(
M +
jω(
−
R
NE
NE
R
+
FE
R
R
M
FE
IND
j ω(
M + M
FE
S
L
+
CAP
NE
R
CAP
FE
m
S
R
)
L
+
)
m
L
R
+
L
R
R
+
NE
NE
R
R
R
+ R
NE
NE
FE
FE
R
+ R
FE
RLC
m
R + R
FE
S
S
L
)
RLC
m
R + R
L
)

Effect of Load Impedance

Common-impedance Common impedance Coupling
^
V
^
V
^
V
^
V
NE
S
FE
S
= jω(
M + M ) +
IND
NE
IND
FE
CAP
NE
= jω(
M + M ) +
CAP
FE
M
M
CI
NE
CI
FE

Time-Domain Time Domain Crosstalk for R=50Ω
R=50

Time-Domain Time Domain Crosstalk for R=1KΩ
R=1K

The Capacitance Equivalent for
the Shielded Receptor Wire

The Lumped Equivalent Circuit for
V
Capacitive Coupling
R
^ CAP ^ CAP
NE FE RS GS
NE = V FE ≅
jω
VG
DC
RNE
+ RFE
C RS + CGS
R
C
C

Illustration of Placing a Shield
on Inductive Coupling

The Lumped Equivalent Circuit
^
V
IND
NE
=
for Inductive Coupling
R
NE
R
+
NE
R
FE
jωL
GR
^
I
G
R
SH
R
+
SH
jωL
SH
SF
=
R
SH
R
+
SH
jωL
SH

Explanation of the Effect
of Shield Grounding

Twisted Wires

The Inductive-Capacitive
Inductive Capacitive
Coupling Model

Terminating a Twisted Pair

A Model for the Unbalanced
Twisted Receptor Wire Pair

Explanation of the Effect
of an Unbalanced Twisted Pair

The Three Levels of
Reducing Inductive Crosstalk

A Coupling Model
for the Balanced Termination

The Effect of Balanced
and Unbalanced Terminations

Purposes of a Shield
� To prevent the emissions of the electronics
of the product from radiating outside the
boundaries of the product
� To prevent radiated emissions external to
the product from coupling to the product’s
electronics

Degradation of Shielding
Effectiveness

Termination of a Cable Shield
� The cable shield may become a monopole antenna, if
the ground potential is varying
� Peripheral cables such as printer cables for PC tend
to have lengths of order 1.5m, which is a quarter-
wavelength at 50MHz
to a Noisy Point
� Resonances in the radiated emissions of a product due
to common-mode currents on these types of
peripheral cables are frequently observed in the
frequency range of 50-100MHz

dB
Shielding Effectiveness
SE = R + A +
dB
dB
M
dB
� R represents
the reflection loss
� A represents
the absorption loss
� M represents
the additional effects
of multiple reflections
/ transmissions

R
dB
≅
Reflection Loss
20
log
10
� By referring to
copper,
R
dB
=
168
+
10
ηo
( )
4η
log
10
≅
(
20
σ
μ
log
� The reflection loss is larger at lower
r
r
f
)
10
(
1
4
σ
ωμ
frequencies and high-conductivity metals
r
ε
o
)

Absorption Loss
t / δ
AdB = 20 log 10 e = 131 . 4t
fμ
rσ
r
� The absorption loss increases with increasing
frequencies as f

Shielding Effectiveness

Shielding Effectiveness
� Reflection loss is the primary contributor to
the shielding effectiveness at low frequencies
� At the higher frequencies, ferrous materials
increase the absorption loss and the total
shielding effectiveness

Shielding Effectiveness of Metals

The Methods of Shielding against
Low-Frequency Low Frequency Magnetic Fields
� The permeability of ferromagnetic materials decreases
with increasing frequency
� The permeability of ferromagnetic materials decrease
with increasing magnetic field strength

The Frequency Dependence
of Various Ferromagnetic Materials

The Phenomenon of Saturation of
Ferromagnetic Materials

The Bands to Reduced the
Magnetic Field of Transformer
Leakage Flux

Effects of Apertures
Since it is not feasible to determine the direction of the
induced current and place the slot direction appropriately,
a large number of small holes are used instead

ESD Events
� Typical rise times are of order 200ps-70ns, with a
total duration of around 100ns-2μs
� The peak levels may approach tens of amps for a
voltage difference of 10kV
� The spectral content of the arc may have large
amplitudes, and can extend well into the GHz
frequency range

Effects of the ESD Events
� The intense electrostatic field created by
the charge separation prior to the ESD arc
� The intense arc discharge current

Three Techniques for Preventing
Problems Caused by an ESD Event
� Prevent occurrence of the ESD event
� Prevent or reduce the coupling (conduction or radiation)
to the electronic circuitry of the product (hardware
immunity)
� Create an inherent immunity to the ESD event in the
electronic circuitry through software (software
immunity)

Preventing the ESD Event
� Electronic components such as ICs are placed in pink
polyethlene bags or have their pins inserted in antistatic
foam for transport
� Some products can utilize charge generation prevention
techniques
� For example, printers constantly roll paper around a
rubber platen. This causes charge to be stripped off the
paper, resulting in a building of static charge on the rubber
platen.
� Wires brushes contacting the paper or passive ionizers
prevent this charge building

Hardware Immunity
� Secondary arc discharges
� Direct conduction
� Electric field (Capacitive) coupling
� Magnetic field (Inductive) coupling

Preventing the Secondary
Arc Discharges

Single-point Single point Ground

Use of Shielded Cables to
Exclude ESD Coupling

The Methods of Preventing
ESD-induced ESD induced Currents

Reduction of Loop Area in
Power Distribution Circuits

Reduction of Loop Areas to Reduce
the Pickup of Signal Lines

Software Immunity
� Watchdog routines that periodically check
whether program flow is correct
� The use of parity bits, checksums and errorcorrecting
codes can prevent the recording of
ESD-corrupted data
� Unused module inputs should be tied to ground
or +5V to prevent false triggering by an ESD
event

Packaging Consideration
� A critical aspect of incorporating good EMC design is
an awareness of these nonideal effects throughout
the functional design process
� Another critical aspect in successful EMC design of a
system is to not place reliance on “brute force fixes”
such as “shielding” and “grounding”

Common-impedance Common impedance Coupling

The Effect of Conductor
Inductance on Ground Voltage

Segregation of Grounds

Ground Problems between
Analog and Digital Grounds

The Generation and Blocking of
CM Currents on Interconnect Cables

Methods for Decoupling
Subsystems

Interconnection and
Number of PCBs
� It is preferable to have only one system PCB rather
than several smaller PCBs interconnected by cables
� The PCBs can be interconnected by plugging their
edge connectors into the motherboard

Use of Interspersed Grounds
to Reduce Loop Areas

PCB and Subsystem Placement
Attention should be paid to the placement and
orientation
of the PCBs in the system

Decoupling Subsystems
� Common-mode currents flowing between subsystems can
be effectively blocked with ferrite, common-mode
chokes
� Another method of decoupling subsystems is insert a
filter in the connection wires or lands between the
subsystems. This filter can be in the form of R-C packs,
ferrite beads, or a combination
� High-frequency signals on the power distribution system
between subsystems can be reduced by the use of
decoupling capacitors

Splitting Crystal/ Oscillator Frequencies
� The 16 th harmonics (32MHz and 31.696MHz) are separated by
304kHz, so that they will not add in the bandwidth of the receiver
� The 100 th harmonic of the 2MHz signal (200MHz) and the 101 st
harmonic of the 1.981MHz signal (200.081MHz) will be within
81kHz of each other and will add in the bandwidth of the receiver

Component Placement

Component Placement

A Good Layout for a
Typical Digital System

Creation of a Quiet Ground
where Connectors Enter a PCB

Unintentional Coupling of Signals
between Chip Bonding Wires
� Placing a small inductor in series with that pin to block
the high-frequency signal
� Ferrite beads could also be used, but their impedance is
typically limited to a few hundred ohms

Use of Decoupling Capacitors

Decoupling Capacitor Placement

Minimizing the Loop Area of
the Power Distribution Circuits