Development and testing of a 10 GHz phased-array cylindrical ...

Development and Testing of a 10 GHz
Phased-Array Cylindrical-Antenna
Transmitting System Incorporating a
Least-Squares Radiation-Pattern
Synthesis Technique
Nikolaos c. Athanasopoulos 1 , Konstantinos A. Mourlzoukos 1 , Giorgos E. Stratakos 2 ,
Rodoula J. Makri 1 , and Nikolaos K. Uzunoglu 1
1Microwaves & Fiber Optics Laboratory, School of Electrical & Computers Engineers
National Technical University of Athens
9 Iroon Polytechniou Str. 15773 Zografou, Athens, Greece
Tel: +302107722289; Fax: +302107722291;
E-mail: nathan@esd.ece.ntua.gr.mourtzoukosk@gmail.com. rodia@esd.ece.ntua.gr, nuzu@cc.ece.ntua.gr
2Advanced Microwave Systems Ltd, 2
25th Martiou Str. 1778 Tauros, Athens, Greece
Tel: +30 2104838442; E-mail: y.stratakos@ams-mw.com,
Abstract
This research presents the development and testing procedures for an experimental phased-array cylindrical-antenna
transmitting system, operating at 10 GHz. The antenna-element excitation phases are set at the intermediate-frequency
stage, and are determined by using a least-squares radiation-pattern synthesis technique. The antenna-element excitation
amplitudes are taken to be equal and fixed. The system provides steerable main lobes and nulls at predefined directions,
including control of the sidelobes at specified levels. It can be used for communications and radar applications with
interference-rejection requirements.
Keywords: Phased arrays; cylindrical arrays; least squares methods; antenna radiation patterns; pattern synthesis; beam
steering; array signal processing
T he
1. Introduction
idea of developing antenna array systems with electronically
controlled beams was proposed as early as the 1940s [1­
4]. In the last three decades, with the advances in MMIC technology,
phased-array systems have started to become feasible at a reasonable
cost. This is illustrated by the many solid-state active
arrays that use MMIC technology that have been deployed or are
under development [5]. The use of phased-array technologies provides
significant benefits compared with conventional mechanically
steered antennas, such as fast searching of a given field of
view, interference-suppression capability, and flexible control of
radiation patterns [6]. Typical phased arrays make use of large
numbers ofelements (several hundred), and usually suffer interference.
This condition does not hold true for the case ofsmall phased
arrays, such as might be used in communication and radar applications.
In some antenna applications, planar arrays have limitations
imposed primarily by the need for introducing an aerodynamic
radome. If array elements are allocated over a confonnal surface,
those limitations may ease. Cylindrical arrays - the most common
type of confonnal arrays - find applications in situations where
wide scanning angles in azimuth are needed, exploiting their azimuthal
symmetry [7]. This benefit has found applications in broadcast
antennas and direction-finding antenna systems [8].
This work presents the development of a small phased-array
transmitting system of cylindrical geometry. Since the literature
lacks small cylindrical phased arrays, the proposed system can be
considered as a demonstrator. It can be used in short-range communication
and radar applications.
In most practical phased-array systems, RF beamfonning is
implemented with both amplitude and phase control of the excita-
80 ISSN 1045-9243120081$25 ©2008 IEEE IEEE Antennas and Propagation Magazine, Vol. 50, No.6, December 2008

tion currents. In this work, beamforming is implemented at the
intermediate-frequency (IF) stage with phase-only control, providing
a faster and less complicated approach for beam steering and
interference rejection. Nevertheless, IF processing introduces phase
unbalance among RF emitted signals, and this must be compensated
for.
In Section 2, the development of the experimental phased
array is presented. The antenna array and the IF and RF units are
given in detail. In Section 3, the radiation-pattern-synthesis method
used is given. It is based on a least-squares technique adapted to
the system's architecture. In Section 4, excitation amplitudes are
adjusted and the phase unbalance among RF emitted signals is
compensated for. Finally, in Section 5, radiation-pattern measurements
are presented for five beamforming scenarios, and these are _
l----~----~-----------:-~~j~~-----~----------~-----][~~~~~~~;~~~~-] -~~~;~:~~~]
, I I I
I I I I
: : : e1.#-10 :
I I,
I
I : TxRx#-5 e1.#-9! I
~------ -- --- --- -------------- -------- ---- ~ :--------------- ------
compared with simulated results. The cylindrical array gain is cal- Figure 2. A conceptual model of the experimental phased-array
culated with the application ofthe Friis transmission equation. antenna system.
2. The Phased-Array
Cylindrical Antenna System
The experimental phased-array antenna system is presented in
Figure 1. As depicted in the conceptual model of Figure 2, the
system was developed in three basic units: the beamforming unit,
the transceivers unit, and the antenna-array unit. The intermediate-frequency
input signal is divided into 11 individual IF signals
that are phase adjusted through the beamforming unit. These 11 IF
phase-adjusted signals are driven in parallel to the 11 transmitter
modules (numbered Tx-5 to Tx5) of the transceivers unit, they
follow up-conversion and amplification procedures, and finally
they are emitted by the 21-element (numbered from element -10 to
element 10) antenna-array unit at the 10 GHz frequency region.
Each pair ofadjacent radiators is fed by the same transmitter module,
with the exception of the central module (element 0), which is
driven alone by the TxO module. In the subsections that follow, the
individual units ofthe system are described.
2.1 The Antenna-Array Unit
The antenna-array unit, originally designed in the framework
of the Western European Armament Group (WEAG)-THALES
JP1.03-WEAG program [9-11], is a conformal array antenna con-
Figure 3. The antenna-array unit.
el.#-2
el.#-1
e1.#O
e1.# 1
e1.#2
Figure 4. A pair of adjacent radiating elements.
el.#9
el.#10
Ground
plane
sisting ofa series of21 microstrip-patch elements. It is printed on a
dielectric substrate of Teflon R04003 (G r = 3.38, tan l5 = 0.0018,
thickness 0.51 mm). The whole structure is mounted on a cylindrical
aluminum surface (with a radius of curvature of 0.5 m). The
antenna-array unit is presented in Figure 3.
The 21 microstrip elements are of rectangular shape, with
dimensions L = 7 mm and W = 11 mm. The center-to-center spacing
between adjacent radiators is 0.66...1, (where A is the free-space
wavelength at the 10 GHz operating frequency), which corresponds
to a 2.25° angular spacing. Each individual radiator is fed through
a coaxial probe, which runs through the aluminum surface and the
dielectric substrate, and finally touches the surface of the element.
Figure 4 presents a pair ofadjacent radiators. The return loss at the
input of each radiator is, on average, -15 dB. The mutual coupling
Figure 1. The experimental phased-array antenna system. between adjacent radiators when they are excited in-phase is, on
fel::l:: Antennas ana propagation Magazine, Vol. 50, No.6, December 2008 81
I
I

average, -24 dB; for the nonadjacent radiators, this value is significantly
reduced.
2.2 The Transceivers Unit
The transceivers unit consists of 11 modules. After being
phase adjusted through the beamforming unit, the 11 IF signals are
introduced in parallel to the transmitting chains of the modules at
140 MHz. After being filtered and up-converted at 1 GHz, they
pass through the appropriate filters and are then amplified. The
final up-conversion is at 10 GHz, where the signals are filtered and
amplified. Each one of the 10 GHz amplified signals is divided by
a power splitter, and drives two adjacent radiators of the 21-element
antenna array_ This means that each module drives two adjacent
radiators, with the exception of the central radiator (element
0), setting the reference phase of the system. This reference element
is driven by a single module (TxO), as is depicted in the conceptual
model of Figure 2. A picture of a transceiver module is
shown in Figure 5.
2.3 The Beamforming Unit
The beamforming unit is implemented at the IF stage. In a
personal computer (PC), the operator sets the system's operation
mode (scanning or non-scanning), defines the number of active
radiators of the 21-element array antenna, and finally specifies the
desired radiation-pattern characteristics, i.e., the main-lobe and null
directions. The operator's environment is presented in Figure 6.
The radiation-pattern specifications that the operator defines
are used to compute the appropriate phase-shift values that must be
set for the 11 IF signals. For that purpose, a least-squares method is
implemented, and it will be presented analytically in the next section.
These values are transferred in digital form through an RS232
serial interface to the control subunit, where they are organized in
12-bit digital sequences while the appropriate control and synchronization
signals are generated. The 12-bit digital sequences are
driven to the digital-to-analog (D-to-A) conversion subunit. There,
under the guidance ofcontrol and synchronization signals, they are
converted to analog voltages in the range of 0-12 V. These analog
Figure 5. A transceiver module of the transceivers unit.
~~~:~.~~·;iC u .
,~;;..~", No"-al~;:; q, ;,:~ rif~:-,',:~,',:,;'~~,'r>
,-.,;;;~.................·:i; :.":'''' :':/11'
.111 _... ..... ;"=·~:;·r::~
";of~ S;i..~f~: ;;;~~~1J-"----
:[:~21
.};~:4t ~.~);: il;l.il \f~;:· ;;,:: ~;. ;~iHEHW
/'1 I ~t~>-, l": 'r, P':\"T
~~~
):;"'i~~ ~ 3J~/~'" /'::t 11,,/:)
Figure 6. The operator environment.
Figure 7. The phase-shifting subunit.
• .• ~ ;:"X~.~
:.. '.;.;::....:;:::'
'I ~:~~~~ :~:~~;=:~sed el96008N1 ~·i;.i:·.':':
~,:;.,;.i'j
voltages are finally driven to the phase-shifting subunit, presented
in Figure 7. It consists of 11 channels, one for each IF signal. Two
phase shifters, controlled by the analog voltages in the range of 0­
12 V, are connected in series at each IF channel. This covers a total
range of -180 0 ~ 180 0 , and sets the appropriate phase shift for
each IF path.
3. Phase-Computation and
Radiation-Pattern Synthesis Technique
The power intensity at the azimuth plane of this 21-element
cylindrical array is expressed in the form [12]
U(¢)=BI:~:f(¢-n~¢)la21
eXP{J 2:R [cos(¢-n~¢)J-La2t (1)
B is a normalization constant, ~t/J is the angular distance between
adjacent elements, R is the radius of the cylinder, and A is the
wavelength at the frequency of operation. In order to achieve a
desired radiation pattern, the nth radiator's phase excitation, La~,
must be appropriately set. In Equation (1), f(t/J-n~t/J) is the
82 IEEE Antennas and Propagation Magazine, Vol. 50, No.6, December 2008

antenna-element pattern, which, for the rectangular microstrip
patch element, is given approximately by [13]
f(¢ -nl1¢) = Ccos(¢-nl1¢)
. [kh cos(¢ - nl1¢)] . [kL sin(¢ - nl1¢)]
SIn SIn
2 2
khcos(¢-nl1¢) kLsin(¢-nl1¢)
2 2
(2)
C is a normalization constant, h is the dielectric substrate thickness,
L is the microstrip radiator's length, and k is the wavelength
number.
A least-squares radiation pattern-synthesis technique [14] is
implemented for the appropriate phase computation. According to
this, assume a required power-intensity pattern mask, M (¢), the
shape of which is specified based upon the requirements of the
main-lobe orientation, the null angles, and the sidelobe levels.
Then, a vector ofphase excitations,
is computed. This will produce a normalized power-intensity pattern,
U0 (¢), in a such way as to minimize the discretized mean-
square difference:
n 2
e 2 = Lluo(l1Jj,(W))-M (l1Jj)1 '
;=1
with n sufficiently large. According to the system architecture, the
excitation phases must be equal for every two adjacent radiators,
with the exception of the central radiator. In other words, the following
constraint must be satisfied:
o 0
La-IO = La_9,
The power-intensity pattern-mask function used is described by the
following equation:
¢o is the desired orientation of the main beam, and ¢I is the
desired position of the null. G is the desired null depth, while D is
the desired sidelobe level. l1¢o and 11¢I set the width of the main
bean and null, respectively.
O,¢ E [¢O - t!,.¢o ~ ¢ ~ ¢o + l1¢ol)
M(¢) = G,l1J E [¢I -11¢I ~ ¢ ~ ¢I + 11¢I] .
{
D,¢ E elsewhere
This radiation-pattern synthesis technique and the computation
of the excitation phases for the corresponding beamforming
scenarios were executed using MathCAD 9 software.
(4)
(5)
(6)
4. Calibration Procedures
4.1 Amplitude Adjustments
The beamforming unit computes and controls only the phases
of the excitation currents to achieve the desired radiation patterns.
This presupposes that amplitudes should be constant. As can be
seen from Figure 8, amplitude variations do not exceed ±0.75 dB
around 11.75 dBm, and, as a result, they can be considered to be
constant.
4.2 Phase Calibration
In each transmission channel, the 140 MHz input signal goes
through power divisions, amplifications, and up-conversions, as
well as suffering from different cable lengths. All these unavoidable
factors introduce phase offsets among the 21 signals that are
finally emitted at the 10 GHz region. This phase offset must be
measured and taken into account in the beamforming procedure, so
that the signals have their phase adjusted correctly.
Phase calibration is implemented by the use of a "Magic T."
Based on the basic properties of the Magic T [15], when two inphase
signals are inserted into Ports 3 and 4, respectively, the
Port 1 output signal is maximized. Simultaneously, the Port 2 output
signal is minimized. The system's phase reference (the output
ofTxO) is connected to Port 3. The Port 1 output drives a spectrum
analyzer, and Port 2 is terminated. The outputs ofthe rest ofthe 10
transmitter modules are routed successively to Port 4. For each of
them, the control voltages of the corresponding phase shifters are
manually progressively changed until the power in the spectrum
analyzer becomes maximum. Port I is then terminated and Port 2
is input to the spectrum analyzer. If the value is minimum, the two
signals in Ports 3 and 4 are in-phase. Analog voltages are translated
to the corresponding phase values, and the phase offsets between
the system's reference and each ofthe inputs ofthe rest ofradiating
elements are calculated. The measured phase offset is presented in
Figure 9. It was observed that the phase offset was the same for
every two adjacent radiators. This was due to the fact they were
driven by the same transmitter module.
E l-------tatr---+-~~_+_---llto.............-.4~~--~
en ~ l----~/!---4--+-~H-_t_-------T---
CD
"C
.a ~-----~---lL....--~H-t----------'t-------
=a:
E~
, -._..,-~-~~~ ·_· ·1~·~ · · · ·· ··..· ·..·· _.__ -:
« J...------+-+-~_4+1t+--------~T:;~
-10 -9 -8 -7 -6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6 7 8 9 10
Number of antenna element
Figure 8. The amplitudes of the signals at the inputs of the 21
radiators.
IEEE Antennas and Propagation Magazine, Vol. 50, No.6, December 2008 83

-~---"-_.._._-_._--_ _ !t..,.._ _ _.---'---'---"---' ;
·8 ·7 -6 -5 -4 -3 ·2 -1
.........•..._..&..0# _ , _ _.- _ _ .......--.
Number of radiator
Figure 9. The phase offset between 10 GHz signals at the
inputs of the radiators and the system's reference.
5. Radiation Measurements
5.1 Radiation Patterns
The phase excitations of the 21 signals that drive the array
antenna, and that are calculated following the least-squares synthesis
technique, are the sum oftwo factors:
T T T
W = WADJ +W CAL '
WIDJ is the vector of phases that must be adjusted by the beam­
forming unit, and WtAL is the vector of the measured phase offset
that is provided by Figure 9. The phases that must be set by the
beamforming unit are then given by
T T T
WADJ = W - WCAL .
The phased-array transmitting system that was developed was
tested through several beamforming scenarios. An X-band hom
antenna (20 dB gain), with its output going into a spectrum analyzer,
was used as a receiver. The distance between the transmitting
system and the receiver was 15.5 m, satisfying the far-field
condition. In the following, azimuth-plane radiation-pattern measurements
are presented for five beamfonning scenarios, and are
compared with simulated patterns. In the diagrams, the red lines
correspond to the measurements, whereas the blue lines correspond
to the simulation. For each scenario examined, the calculated phase
excitations, W T , are given in the corresponding tables (Tables 1 to
5).
Table 1. The calculated antenna-element excitation phases with
the main lobe at 0° and SLLs < -20 dB.
84
# ofRadiator W # ofRadiator W
(7)
(8)
-10 -4.469° 1 -12.319°
-9 -4.469° 2 -12.319°
-8 139.812° 3 -47.957°
-7 139.812° 4 -47.957°
-6 -115.279° 5 -116.081°
-5 -115.279° 6 -116.081°
-4 -47.154° 7 140.042°
-3 -47.154° 8 140.042°
-2 -11.975° 9 -2.578°
-1 -11.975° 10 -2.578°
0 0° - -
Table 2. The calculated antenna-element excitation phases with
the main lobe at 10°, SLLs < -20 dB.
# of Radiator W # of Radiator W
-10 -51.108° 1 39.591°
-9 -51.108° 2 39.591 °
-8 178.018° 3 78.839°
-7 178.018° 4 78.839°
-6 9.339° 5 85.027°
-5 9.339° 6 85.027°
-4 168.564° 7 50.535°
-3 168.564° 8 50.535°
-2 -68.64° 9 -20.512°
-1 -68.64° 10 -20.512°
0 0° - -
Table 3. The calculated antenna-element excitation phases with
the main lobe at 20°, SLLs < -20 dB.
# of Radiator W # ofRadiator W
-10 -64.973° 1 103.773°
-9 -64.973° 2 103.773°
-8 -116.425° 3 -144.385°
-7 -116.425° 4 -144.385°
-6 158.892° 5 -62.395°
-5 158.892° 6 -62.395°
-4 35.924° 7 -15.355°
-3 35.924° 8 -15.355°
-2 -122.498° 9 5.099°
-1 -122.498° 10 5.099°
0 0° - -
Table 4. The calculated antenna-element excitation phases with
the main lobe at 0°, a null at 20°, SLLs < -20 dB.
# ofRadiator W # ofRadiator W
-10 -45.149° 1 -19.022°
-9 -45.149° 2 -19.022°
-8 107.497° 3 -44.863°
-7 107.497° 4 -44.863°
-6 -140.661° 5 -132.582°
-5 -140.661° 6 -132.582°
-4 -70.932° 7 140.443°
-3 -70.932° 8 140.443°
-2 -19.423° 9 -6.875°
-1 -19.423° 10 --6.875°
0 0° - -
IEEE Antennas and Propagation Magazine, Vol. SO, No.6, December 2008

- 0 -25 -20 -15 -10
ph( degrees)
10 15 20 25 3
ccl-------.,~---J.-+-::-Hr-+--~~----:~------t
~ :::>1--------."J---~+_____::_9A__+_--_T_J~--_\_--7'i
Figure 10. The main io6e at Oo~
iii'
~~--'--~~~:""'---~i-+-----+-A-+-+-~----:::!i~
::>~:I:::::...--~,------.::w:~------';~-+--I-----i
phi (degrees)
mJ-----P:---f.S--\---~Ir4_--__4_I~_l_----~
~
::>I-----~+----lt~__+_~~--_+_~_;_----_i
phi (degees)
Figure 12. The main lobe at 20°.
phi (degrees)
Figure 11. The main lobe at 10°.
-25 -20 -15 -10 -5
15 18 21 24 27 3D
in L...J.,.--------...-2QL.+----~t:=::~--_____1
"'0
......" 1---~-___J~~--=~ri5-+-----+_-_::__-____j
-----_._--~~---phi
(degrees)
Figure 13. The main lobe at 0° degrees with a null at 20°. Figure 14. The main lobe at 10° with a null at 16°.
IEEE Antennas and Propagation Magazine, Vol. 50, No.6, December 2008 85

Table 5. The calculated antenna-element excitation phases with
the main lobe at 10°, a null at 16°, SLLs < -15 dB.
# of Radiator W # of Radiator W
-10 -53.113° 1 50.821 °
-9 -53.113° 2 50.821°
-8 145.589° 3 99.981 °
-7 145.589° 4 99.981 °
-6 -19.652° 5 106.284°
-5 -19.652° 6 106.284°
-4 152.006° 7 63.369°
-3 152.006° 8 63.369°
-2 -75.401 ° 9 -18.449°
-1 -75.401° 10 -18.449°
0 0° - -
5.1.1 Main Lobe at 0°, SLLs < -20 dB
The calculated phase excitations for this case are given in
Table 1. It was observed from Figure 10 that the measured pattern
had its maximum in the frontal direction. The sidelobe levels
(SLLs) varied below -15 dB, while the specified levels, having
been fulfilled by the simulated pattern, were -20 dB.
5.1.2 Main Lobe at 10°, SLLs < -20 dB
The calculated phase excitations for this case are given in
Table 2. Figure 11 shows that the main lobe was steered at 10°.
Two sidelobes were at about -10 dB, while the rest ofthe sidelobes
varied around -15 dB. The specified sidelobe levels, having been
fulfilled by the simulated pattern, were -20 dB.
5.1.3 Main Lobe at 20°, SLLs < -20 dB
The calculated phase excitations for this case are given in
Table 3. The main lobe was focused at 20°, as Figure 12 depicts.
The sidelobes varied around -15 dB, while the specified levels
were -20 dB. In the region around -20°, a sidelobe with a level of
-5 dB appeared, as the simulation predicted.
5.1.4 Main lobe at 0°, Null at 20°,
SLLs < -20 dB
The calculated phase excitations for this case are given in
Table 4. Figure 13 shows that the main lobe was focused on the
frontal direction. A null was placed at 20°, as asked. Its depth was
about -28 dB, while the specified value, having been fulfilled by
the simulated pattern, was -50 dB. The sidelobes varied around the
-15 dB level, while the specified levels, having been fulfilled by
the simulated pattern, were -20 dB.
86
5.1.5 Main lobe at 10°, Null at 16°,
SLLs < -15 dB
The calculated phase excitations for this case are given in
Table 5. It was observed from Figure 14 that the main lobe was
steered to 10°, and a null was placed at the requested position, 16°.
Its depth was about -21 dB, while the specification, having been
fulfilled by the simulated pattern, was -20 dB. The sidelobes varied
around -15 dB, as was specified and was fulfilled by the simulated
pattern.
5.2 Gain Calculation
The gain of this 21-element cylindrical array was computed
using the Friis transmission equation. According to that equation,
the received power that is measured in the spectrum analyzer is
given by [19]
P,. =GARRAr GHORN~ (l-IrARRAl)(l-IrHORN 1 2 )
(4~S )\Vt • PHORNIL. (9)
The gain ofthe array in dB is then given by
[ n=lO ( 2)]
(GARRAr)dB=(P")dBm-lOlog n~o~nX l-Irt.l
-lOIOg(l-IrHORNn-(GHORN )dB - 2010g( 4~S)
-IOlog(lpt· PHORN/)-(L)dB (10)
The power received, p,., in the spectrum analyzer was measured as
-20 dBm. F;n (n = -10,...,10) is the power that was fed to the 21
radiators and arises from Figure 8. r t (n =-10,...,10) is the
n
return loss at the input of each radiator, which, as was already
mentioned in Section 2.1, was -15 dB. r HORN is the return loss at
the input of the hom antenna, which was -8 dB, while, as has
already be~n mentioned, the hom antenna's gain was 20 dB. The
free-space losses for the distance 8 = 15.5 m and for A = 3cm are
20 10g(~) = -76.25 dB.
41£8
(11)
During measurements, it was observed that there were no crosspolarization
phenomena between the 21-element cylindrical array
and the hom antenna. As a result of that, the factor Ipt· PHORNI
could be ignored in Equation (10). Finally, in Equation (10) the
factor L represents the losses of the coaxial cable that carries the
received signal from the hom antenna to the spectrum analyzer.
These losses were measured as being -3 dB.
Taking into account all the factors involved in Equation (10),
the 21-element cylindrical array gain can be computed as
IEEE Antennas and Propagation Magazine, Vol. 50, No.6, December 2008

(GARRAy);;LINDRlCAL = 14.25 dB.
Assuming now that these 21 radiators were allocated over a planar
surface, the maximum planar array gain would ideally be 21 times
the element gain [1 7]. The element gain was measured to be
5.5 dBi, so for this ideal planar-array model, the gain would be
PLANAR ()
( GARRAy ) dB =lOlog 21 +5.5=18.72dB.
(12)
(13)
Equation (13), which gives the gain for the planar array, is
ideal. If a realistic planar array was tested by application of the
Friis transmission formula, the array gain would be lower than the
ideal value of 18.72 dB (Equation (13)). A quantitative comparison
of the planar and cylindrical array gain would make sense only if
both arrays were measured with the same method, and the Friis
transmission formula was applied. The purpose of the reference to
the planar case in the framework ofthis paper is only qualitative, in
order to show that the radiators that approach the edges ofthe array
contribute less than the element's maximum power towards the
array's frontal direction.
Finally, from the patterns ofFigures 10-12, it can be observed
that the -3 dB beamwidth was 5°, staying constant as the main
beam scanned. Consequently, the cylindrical-antenna gain did not
fluctuate within the -20° ~ 20° scanning area.
5.3 Discussion
The measured radiation patterns proved that this cylindrical
phased-array antenna system provides steerable main beams at prespecified
~irections, and nulls at desired positions, of the azimuth
plane. The IF phase control, applying a least-squares pattern-synthesis
technique in combination with phase calibration, gave accurate
results in terms ofthe main-beam and null positions. This was
shown by the good agreement between measurements and simulations.
On the other hand, it was observed that the sidelobe levels
were substantially higher than the desired specifications, about
5 dB to 10 dB above the specifications. The small variation in the
amplitudes of the emitted signals (as presented in Figure 8), may
have been a possible reason for this. The mutual coupling among
the radiating elements, which varied as the phase excitations
changed, may also have affected the radiation characteristics in
terms ofthe sidelobe levels.
As mentioned in the introduction, cylindrical arrays find
application in situations where wide scanning angles in azimuth are
needed, exploiting their azimuthal symmetry. In this research, as
the inter-element angular spacing was 2.25°, the 21-element array
covered a 45° arc of the cylindrical surface, illuminating only a
sector of the azimuthal plane. For the total azimuthal illumination,
eight such 21-element arrays could be successively allocated on the
cylindrical surface, and they would create a full ring array, which
could be fed by the proposed system. An appropriate commutation
network, that could be incorporated into the RF stage of the system,
would feed one 21-element subarray each time, illuminating
the corresponding azimuthal sector while the seven remaining 21element
subarrays stayed inactive.
IEEE Antennas and Propagation MagaZine, Vol. 50, No.6, December 2008
6. Conclusions
The development of a 10 GHz cylindrical-array antenna
transmitting system was presented in this study. The system was
measured and tested, and the radiation patterns achieved were
compared with the simulated patterns. Good agreement in the
comparison showed that the use ofIF beamforming proved to be an
approach providing stability and easy control of the radiation patterns.
The system provides radiation patterns with steerable main
lobes, and nulls at prespecified positions within the azimuthal
region of -20° ~ 20° . The scanning speed ofthe system is 8 JLs. It
can be used in radar and communication applications with interference-rejection
requirements. Additionally, its basic properties can
be applied in MIMO communication systems [18, 19]. A MIMO
system has multiple antenna elements at both the transmitter and
receiver. Those multiple antennas can be used for the transmission
of several parallel data streams to increase the capacity of the system.
This presupposes the ability to control the excitation of the
multiple antennas for both the transmitter and the receiver. The
phased controlled system that was presented in this research gives
the ability to apply such technology towards confronting non-desirable
effects caused by multipath propagation. Finally, the possibility
ofusing the developed conformal array as a focal-plane radiator
in a large reflector antenna is an interesting application that is also
being considered.
7. References
1. M. I. Skolnik, "Survey of Phased Array Accomplishments and
Requirements for Navy Ships," Phased Array Antennas, Norwood,
MA, Artech House Inc., 1972, pp. 15-20.
2. L. N. Ridenour, Radar System Engineering, New York,
McGraw-Hill Book Co., 1947, pp. 219-295.
3. R. A. Smith, Aerials for Metre and Decimetre Wave-Lengths,
Cambridge, Cambridge University Press, 1949.
4. A. Price, Instruments of Darkness, New York, Charles
Scribner's Sons, 1970.
5. E. Brookner, "Phased Arrays Around the World - Progress and
Future Trends," IEEE 2003 International Symposium on Phased
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(ed.), Advanced Signal Processing Handbook, First Edition, Boca
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7. M. I. Skolnik, Introduction to Radar Systems, Second Edition,
New York, McGraw-Hill International Editions, 1981, Chapter 8,
Section 8.10, pp. 330-331.
8. R. J. Mailloux, Phased Array Antenna Handbook, Second Edition,
Norwood, MA, Artech House Publishers, 1994, Chapter 4,
Section 4.2, p. 194.
9. N. K. Uzunoglu, D. 1. Kaklamani, G. E. Stratakos, et aI., "On the
Development of a Conformal Array System," Workshop COST­
260, Dubrovnik, Croatia, December 1997.
87

10. N. K. Uzunoglu, D. I. Kaklamani, G. E. Stratakos, et aI.,
"Design ofa Cylindrically Shaped Confonnal Array System," 28th
European Microwave Conference, Amsterdam, Netherlands, 1998.
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Diamantis, Spyridakis, Th. Kaifas, J. N. Sahalos, G. Stratakos, P.
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Printed Cylindrical Antenna Array for Radar Applications," URSI
General Assembly, Maastricht, Netherlands, August 2002.
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Norwood, MA, Artech House Publishers, 1996, Chapter 2,
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13. R. Garg, P. Bhartia, I. Bahl and A. Ittipiboon, Microstrip
Antenna Design Handbook, Norwood, MA, Artech House, 2001,
Chapter 4, Section 4.2.2, pp. 257-264.
14. S. Prasad, and R. Charan, "On the Constrained Synthesis of
Array Patterns with Applications to Circular and Arc Arrays,"
IEEE Transactions on Antennas and Propagation, AP-32, 7, July
1984, pp. 725-730.
15. N. Marcuvitz, Waveguide Handbook, New York, Dover Publications,
1965, Chapter 7, Section 3-2.
16. C. Balanis, Antenna Theory Analysis and Design, Second Edition,
New York, John Wiley & Sons, 1997, Chapter 2, Section
2.17.1, pp. 86-88.
17. R. J. Mailloux, Phased Array Antenna Handbook, Second Edition,
Norwood, MA, Artech House, 1994, Chapter 2, Section 2.1.2,
p.68-72.
18. M. A. Jensen and J. W. Wallace, "A Review of Antennas and
Propagation for MIMO Wireless Communications," IEEE Transactions
on Antennas and Propagation, AP-52, II" November
2004 pp. 2810-2824.
19. A. F. Molisch and M. Z. Win, "MIMO Systems with Antenna
Selection," IEEE Microwave Magazine, March 2004, pp. 46-56. 0!)
88 IEEE Antennas and Propagation Magazine, Vol. 50, No.6, December 2008