A Full-Bridge DC-DC Converter with Zero- Voltage Switching ... - ijcee

International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
A Full-Bridge DC-DC Converter with Zero-
Voltage Switching: Experimental Studies
R.samuel Rajesh Babu and Joseph Henry
Abstract—This paper deals with the simulation and
implementation of Full-Bridge DC-DC converter with Zero
Voltage Switching (ZVS). The 48V DC is efficiently reduced to
12V DC using DC-DC converter. Switching losses are
reduced by zero voltage switching. Switching stresses are
reduced by using resonant inductor and capacitor. This
converter has advantages like low switching loss, less EMI, and
less switching stresses. DC-DC converter is modelled and
simulated using simulink. The hardware is fabricated and
tested..The simulation results are compared with the
experimental results. The aim of this work is to develop an
efficient DC-DC converter with high power density.
Index Terms—DC-DC power conversion, soft-switching,
zerovoltage- switching (ZVS).
I. INTRODUCTION
The full-bridge (FB) zero-voltage-switching (ZVS)
converter (FBZVS converter), is the most popular topology
for DC-DC converters due to fixed switching frequency,
ZVS operation, high efficiency, low circulating reactive
energy and moderate device stresses. By using a DC
blocking capacitor and a saturable inductor in series with
primary winding, the primary current during the freewheeling
interval can be reduced to zero. This circuit is
called as the zero-voltage and zero-current switching
(ZVZCS) FB converter wherein the lagging-leg switches
operate at ZCS and leading-leg switches operate with ZVS.
The major limitation of the FBZVS converter has been
the limited range of operation over which ZVS can be
achieved. When the load current is low, the ZVS of the
lagging-leg switches is lost as the energy stored in the
leakage inductance of the transformer is insufficient to
discharge the switch and transformer capacitances. The loss
of ZVS results in increased switching losses and
electromagnetic interference (EMI). In the case of highpower
converters using insulated gate bipolar transistor
(IGBT), an external snubber capacitor is connected to
reduce the rate of rise of voltage and turn-off losses.
Therefore, in high-power converters, the loss of ZVS
additionally results in the discharge of snubber capacitor in
IGBT. The resulting surge current can be detrimental to
IGBT and capacitor in the long run and it increases EMI
problem. Further, the resonant voltage overshoots due to
resonance between the snubber capacitor and wiring/lead
inductance can exceed IGBT voltage rating. Therefore, it is
Manuscript received June 10, 2010.
R.samuel Rajesh Babu, Dr. Joseph henry, Research scholar, EEE
department,Professor, EEE Department. Sathyabama University, Vel Tech
University, Chennai, India.(e-mail: Samuel.rajeshbabu@gmail.com)
important to maintain ZVS operation over the entire range
of operation or the conversion range. The following
solutions have been proposed in the past.
1) Using higher series inductance increases the ZVS range
but results in increased loss of duty cycle and ringing
across secondary-side rectifier diodes. With consequent
reduction in transformer turns ratio, primary reflected
current and switch conduction loss increases.
2) Using saturable inductor instead of a linear inductor,
ZVS range can be increased without significantly losing
the duty ratio. However, a large-size core is required to
implement the saturable inductor.
3) The energy stored in the magnetizing inductance can
also be used to aid the ZVS operation. The switch
current and the conduction loss is significantly increased.
In the converter proposed in and, the stored energy in
the magnetizing inductance of auxiliary transformer
(which is independent of load) is used to extend the ZVS
range.
4) Using a passive auxiliary “pole” circuit, full-range ZVS
operation can be achieved but the fixed circulating
current results in additional conduction loss. In the
above listed techniques, except for, the range of ZVS
operation can be extended at the expense of increased
full-load conduction loss. Ideally, additional energy
storage is not required under full-load condition since
the energy stored in transformer leakage inductance is
sufficient for ZVS operation. The additional stored
energy is required only when the load current is less.
FBZVS converters featuring this kind of adaptive energy
storage using coupled inductors.
II. FB ZVS CONVERTER
The Fig.1 shows the circuit diagram of the proposed
FBZVS converter. Four MOSFET or IGBTswitches, Q1-Q4,
four antiparallel diodes, D1-D4, and four snubber
capacitorsC1-C4, constitute the full-bridge switching circuit.
The differences between the proposed and conventional
FBZVS converter are as follows.
1) The DC blocking capacitor of conventional converter is
split into two capacitors,Cdc1 andCdc2 , in the proposed
circuit.
2) While the conventional converter uses a single highfrequency
transformer, it is divided into two
transformersTr1 and Tr2 (with primary-to-secondary
turns ratio of ) in the proposed circuit.
3) The proposed circuit has additional inductor La which
adaptively stores additional energy for ZVS operation
when the stored energy in transformer leakage is
inadequate.
211

International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
Fig. 1. Circuit diagram of proposed FB ZVS converter with current doubler
rectifier on the secondary side.
switches Q1-Q4, respectively. In steady-state the voltage
across the capacitors Cdc1andCdc2 is equal to (Vd/2) . The
resulting voltage waveforms across the primary windings of
the two transformers are shown as Vp1andVp2 .Due to the
series connection of the secondary windings as shown in
Fig. 1, Vs=(Vp1+Vp2)/N. The waveform of Vs is a threestep
bipolar square-wave voltage waveform with amplitude
equal to ±(Vd/N) and duty cycle D.
The waveform of transformer primary current is shown
as Ip1 and Ip2. The load current is low and the energy
stored in transformer leakage inductance is not sufficient to
itself achieve ZVS of all the switches Q1-Q4. Under this
condition it is desired that the sufficient energy should get
stored in La so that ZVS of switches can be achieved. The
voltage across La can be written as Vla=(Vp1-Vp2) . The
waveform of Vla is a three-step bipolar square-wave voltage
waveform with amplitude equal ± Vd to and duty cycle (1-
D). Therefore, when D is low and load current is less, the
duty cycle of Vla is high. The peak value of Ila is high.
Sufficient energy is thus available in La to achieve the ZVS
operation. Ila is derived as where Fs=(1/Ts) is the switching
frequency.
Fig 2. Transformer primary and secondary side connections for alternative
rectifier configurations for the proposed converter. (a) Full –wave full
bridge rectifier. (b) Full -wave center- tap rectifier.
The secondary windings of the transformers are
connected in series. The leakage inductances of both the
transformers are shown as a lumped inductor Ls in series
with secondary windings. The diodes Dr1,Dr2 inductors
Lf1,Lf2 and capacitor Cf form the output current doubler
rectifier and filter.Ro is the load resistance and Vd is the
input DC voltage source. The current doubler rectifier on
the secondary side in Fig. 1 can be replaced with the fullwave
bridge and center-tap rectifiers if suitable. Primary
and secondary connections of the transformers for
alternative rectifier configurations are shown in Fig. 2.
III. PRINCIPLE OF OPERATION
The idealized waveforms of the converter with proposed
auxiliary circuit in the steady-state are shown in Fig. 3. The
details of switching transitions are not shown explicitly in
the figure because the intension is to describe the operating
principle of adaptive energy storage in the auxiliary
inductor which aids the full-range ZVS operation and the
mechanism of ZVS transitions in FB converters is well
understood. Let D be the duty cycle of the output voltageVs ,
at the terminals of series-connected secondary windings of
transformers Tr1 and Tr2. The key waveforms for the
operation when D is low are shown by the solid dark lines
in Fig. 3.
The voltages Vg1-Vg4 are the gate voltage signals for
The changes in relevant waveforms for the operation of
circuit when D is high are shown by the dashed lines in Fig.
3. The load current is high and energy stored in the
transformer leakage inductance itself is sufficient to achieve
ZVS of the switches. Under this condition it is desired that
the energy storage in La is minimal. It is quite clear from
the above discussion and from the waveforms of Fig. 3 that
the duty cycle of Vla is low. Therefore, Ila is lower and so
is the energy stored in inductor La.
In the applications where output is fixed (e.g. voltage
regulator modules), D is ideally independent of load if
output filter inductor current is continuous. This continuous
conduction mode (CCM) of operation is, however,
practically restricted to typically up to 20% of the maximum
load current otherwise the required value and size of filter
inductor becomes very large. In discontinuous conduction
mode (DCM) D reduces with the load current and at no-load
condition,D≅0. In applications where the output is required
to be adjustable over a wide range and load resistance is
fixed (e.g., an electromagnet power supply), the expression
for load current (neglecting duty ratio loss) can be written as
I o =(V d /2N ro ) D=I o ,max D
Thus, the load current and the auxiliary inductor current
in the proposed circuit vary opposite to each other. When D
is high, load current is high. Energy stored in transformer
leakage inductance is sufficient for ZVS operation.
Auxiliary current is low causing low additional conduction
losses in the devices. When D is low, load current is low
and energy in transformer leakage inductance is insufficient
for ZVS operation. Auxiliary current increases and assists to
achieve ZVS operation. Thus the trade-off between the ZVS
operation and conduction losses is optimally resolved and
full-range ZVS is achieved without significantly increasing
212

International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
full-load conduction losses . Although the proposed
FBZVS converter has two transformers, the combined
ratings of the two transformers is the same as one
transformer in the conventional FBZVS converter.
the rectifier. For this case the fundamental component of the
square wave voltage is used in the ac analysis.
48v DC is converted into high frequency AC using as
inverter. This is stepped down to 12v by using a step down
transformer. Further this is rectified and filtered using LC
filter. Switching losses are reduced by zero voltage
switching. Switching stresses are reduced by using resonant
inductor and capacitor The even harmonics in the output of
the rectifier are filtered using LC filter. Driving pulses are
applied to the IGBT’S in such a way that the pulse width
coincides with the resonant period.
The primary voltage of the high frequency transformer as
shown in (1),
(1)
The magnetizing current i M (t) is
As for the primary winding, the dot end of the secondary
winding is more positive that the non-dot ends. This implies
that diode D1 & D2 is conducting while D3 & D4 is not
conducting. The secondary voltage can be computed as in
(3).
(2)
Fig .3. Idealized steady-state waveforms of proposed converter
The primary voltage of the individual transformer in
proposed converter (±Vd/2, peak) is half as compared to
that in conventional converter (±Vd , peak) Thus the total
volt-ampere rating of two transformer in proposed converter
is the same as single transformer in conventional converter.
In high-power applications two half-rated transformers in
proposed converter can ease thermal management. Similarly,
the worst-case DC voltage(±Vd) that might appear across
two DC blocking capacitors in the proposed converter is the
same as that in the conventional FBZVS converter.
IV.
ANALYSIS AND DESIGN OF FULL BRIDGE DC-DC
CONVERTER
The Full-Bridge DC-DC converter applies a square wave
of voltage to a resonant network. The resonant network has
the effect of filtering the higher harmonic voltages so that,
essentially, a sine wave of current appears at the input to the
resonant circuit (this is true over most of the load range of
interest). This fact allows classical ac analysis techniques to
be used.
The fundamental component of the square wave input
voltage is applied to the resonant network, and the resulting
sine waves of current and voltage in the resonant circuit are
computed using classical AC analysis. For a rectifier with
an inductor output filter, the sine wave voltage at the input
to the rectifier is rectified, and the average value takes to
arrive at the resulting dc output voltage. For a capacitive
output filter, a square wave of voltage appears at the input
to the rectifier while a sine wave of current is injected into
The current flowing into the inductor was calculated by
using (4),
The output voltage across the inductor Lo as shown in
(5),
The output voltage is shown in (6),
The primary voltage of the high frequency transformer as
shown in (7),
The magnetizing current of the high frequency
transformer is as shown in (8),
(3)
(4)
(5)
(6)
(7)
213

International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
In this instance, as at the primary, the dot ends are more
negative than the non-dot ends, which results in (9).
The output voltage across the inductor is shown in (10).
(8)
(9)
(10)
The output current flowing through the inductor is
shown in (11).
(11)
Output power is shown in Fig 5h. It can be seen that the
output is free from ripple.The specifications are as follows:
DESIGN PARAMETER
Input Voltage
L
C 1 =C 2
L f
C f
R
T ON 50 %
T OFF 50 %
T 100 %
Duty Cycle 50 %
Switching Frequency 50 KHz
Transformer Ratio 4:1
Output Voltage
12V
Output Current
4 A
POWER
48 W
DIODE
IN4007
MOSFET
IRF840
RATING
48V
100 micro H
100 micro F
0.01 pico H
2200 micro F
3 Ω
The output voltage across the capacitor Co was
calculated using (12),
(12)
The output voltage can be obtained across the load is
shown in (13),
(13)
The output power of the converter is shown in (14),
(14)
Fig 5a. Circuit diagram
The average input current can be calculated by using
(15)
where I in,av , is the average input current and δ= 0.8.
The efficiency of the converter is shown in (16),
(16)
Fig 5b. Input voltage
V. SIMULATION RESULTS
ZVS DC-DC converter is modelled using the blocks of
Simulink and the results are presented in this section.The
circuit of ZVS DC-DC converter is shown in fig 5a.Input
voltage is shown in Fig 5b.Driving pulse and voltage across
Q3 switch are shown in Fig 5c. Inverter output voltage is
shown in Fig 5d. Transformer winding –1 is voltage shown
in Fig 5e . Transformer winding –2 is voltage shown in Fig
5f .DC output voltage and current are shown in Fig 5g..
214
Fig 5c. Driving pulse and voltage across Q3 switch

International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
Fig 5d .Inverter output voltage
FB ZVS converter is proposed to achieve ZVS over entire
conversion range with minimum additional conduction loss.
The proposed converter does not use auxiliary coupled
inductor or transformer, rather, the main power transformer
is divided into two half-rated transformers and an uncoupled
inductor is used to achieve ZVS over entire conversion
range. The proposed converter has following advantages
like low switching loss ,less EMI, less switching stresses
and high efficiency. It is particularly suitable in applications
where the output is required to be adjustable over a wide
range and load resistance is fixed (e.g. an electromagnet
power supply).
A. Input Voltage vs Output Power
Fig 5e Transformer winding –1 voltage
Fig 5f .Transformer winding –2 voltage
INPUT
VOLTAGE
OUTPUT
POWER(CONVENTIONAL
CONVERTER)
OUTPUT
POWER(PROPOSED
CONVERTER)
220 915 1089
250 1186 1413
280 1492 1775
B. Input Power vs Output Power
Fig 5g. Output voltage and current
Fig 5h. Output power
VI.
COMPARISON OF CONVENTIONAL AND PRPPOSED FULL
BRIDGE DC-DC CONVERTER
The conventional FB ZVS converter uses a transformer
and an uncoupled inductor to achieve ZVS operation over
the entire conversion range . In this paper a new topology of
INPUT
POWER
OUTPUT
POWER(CONVENTIONAL
CONVERTER)
OUTPUT
POWER(PROPOSED
CONVERTER)
2008 915 1089
2108 1186 1413
2378 1492 1775
215

C. Input Voltage vs Efficiency
International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
Fig 7a.Hardware layout of FB ZVS converter
INPUT
VOLTAGE
EFFICIENCY OF
CONVENTIONAL
CONVERTER
EFFICIENCY OF
PROPOSED
CONVERTER
44 71 79
48 74 81
52 76 83
56 77 84
VII. EXPERIMENTAL RESULTS
The hardware for DC-DC converter is fabricated in the
laboratory with resistive load. Pulses required by the IGBT’
s are generated by using a microcontroller. These pulses are
amplified by using a driver amplifier. The hardware
implementation details are shown in Fig. 7a. The hardware
consists of power circuit and microcontroller based control
circuit. The pulses are generated by using the ATMEL
microcontroller 89C2051.
These pulses are amplified using the driver IC IR2110.
Control circuit for generating the driving pulses is shown in
Fig. 7b. The AC input voltage is shown in Fig. 7c. The
pulses generated by the microcontroller are shown in Fig.
7d. The inverter output voltage is shown in Fig. 7e. The
transformer primary side voltage is shown in Fig. 7f. The
transformer secondary side voltage is shown in Fig. 7g.
The DC output voltage is shown in Fig. 7h. The
specifications are as follows:
DESIGN PARAMETER
RATING
Input Voltage
48V
L
26 milli H
C 1 =C 2
2200 micro F,63 V,83 C
L f
7 milli H
C f
47 micro F,63 V,83 C
R
3 Ω
Transformer
500 milli Amp
MOSFET
IRF840
DIODE
IN5408
Duty Cycle 50 %
Switching Frequency 50 KHz
Transformer Ratio 4:1
Output Voltage
12V
Output Current
4 A
Power
48 W
Fig.7b Control circuit for generating the driving pulses
Fig 7c. AC Input Voltage
Fig 7d. Driving Pulses
216

International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
Fig. 7e. Inverter Output Voltage
VIII. CONCLUSION:
A Full-Bridge DC-DC converter with Zero Voltage
Switching (ZVS) is modelled using the blocks of simulink.
Input DC is converted to high frequency AC and it is
stepped down to 12V level. Latter it is rectified using full
wave rectifier. Soft switched DC-DC Converter is analysed ,
simulated and results are presented. Switching losses are
reduced by zero voltage switching. Switching stresses are
reduced by using resonant inductor and capacitor. This
converter has advantages like higher output current, smaller
transformer and reduced filter size. The efficiency is
improved by using soft switching. Power density is
increased by reducing the volume. The volume is reduced
due to the increased frequency. This converter can be used
for battery charging, control of DC drives Electrolysis and
high power applications. ZVS full bridge DC to DC
converter is simulated and implemented. From the
simulation and experimental results,it can be seen that the
experimental results are similar to simulation results.
Fig 7f. Transformer primary side voltage
Fig 7g. Transformer secondary side voltage
Fig. 7h .DC Output Voltage
REFERENCES
[1] R. A. Fischer, R. D. T. Ngo, and M. H. Kuo, “A 500 kHz, 250W dc–
dcconverter with multiple outputs controlled by phase-shifted PWM
and magnetic,” in Proc. High Freq Power Conv., May 1988, pp. 100–
110.
[2] J. A. Sabate, V. Vlatkovic, R. B. Ridley, F. C. Lee, and B. H.
Cho,“Design considerations for high-voltage high-power full-bridge
zerovoltage- switching PWM converter,” in Proc. IEEE Appl. Power
Electron.Conf. (APEC), 1990, pp. 275–284.
[3] D. Dalal, “A 500 kHz multi-output converter with zero voltage
switching,” in Proc. IEEE Appl. Power Electron. Conf. (APEC), 1990,
pp. 265–274.
[4] F.-S. Tsai, “Small-signal and transient analysis of a zerovoltageswitched,phase
controlled PWM converter using averaged
switch model,” IEEE Trans. Ind. Appl., vol. 29, no. 3, pp. 493–499,
May/Jun.1993.
[5] A. W. Lotfi, Q. Chen, and F. C. Lee, “Non-linear optimization tool
for the full-bridge zero-voltage-switched dc–dc convertor,” Proc.
Inst.Elect. Eng. B, vol. 140, no. 5, pp. 289–296, Sep. 1993.
[6] J. G. Cho, J. A. Sabate, G. Hua, and F. C. Lee, “Zero-voltage and
zero-current-switching full bridge PWM converter for high power
applications,”IEEE Trans. Power Electron., vol. 11, no. 4, pp. 622–
627, Jul. 1996.
[7] G. Hua, F. C. Lee, and M. M. Jovanovic, “An improved full-bridge
zero-voltage-switched PWM converter using a saturable inductor,”
IEEE Trans. Power Electron., vol. 8, no. 4, pp. 530–534, Oct. 1993.
[8] S. Hamada and M. Nakaoka, “Analysis and design of a saturable
reactor assisted soft-switching full-bridge dc–dc converter,” IEEE
Trans.Power Electron., vol. 9, no. 3, pp. 309–317, May 1994.
[9] [R.Watson and F. C. Lee, “Analysis, design and experimental results
of a 1-kW FB-ZVS-PWM converter employing magamp secondary
side control,” IEEE Trans. Ind. Electron., vol. 45, no. 5, pp. 806–814,
Oct. 1998.
[10] R. Ayyanar and N. Mohan, “Novel soft-switched dc–dc converter
with full ZVS-range and reduced filter requirement—Part I:
Regulated-output applications,” IEEE Trans. Power. Electron., vol.
16, no.2, pp. 184–192, Mar. 2001.
[11] R. Ayyanar and N. Mohan, “Novel soft-switched dc–dc converter
with full ZVS-range and reduced filter requirement Part II: Constantinput,
variable-output applications,” IEEE Trans. Power. Electron.,
vol. 16, no. 2, pp. 193–200, Mar. 2001.
[12] P. K. Jain, W. Kang, H. Soin, and Y. Xi, “Analysis and design
considerations of a load and line independent zero voltage switching
full bridge DC/DC converter topology,” IEEE Trans. Power Electron.,
vol. 17, no. 5, pp. 649–657, Sep. 2002.
[13] Y. Jang, M. M. Jovanovic, and Y. Chang, “A new ZVS-PWM
fullbridge converter,” IEEE Trans. Power Electron., vol. 18, no. 5,
pp.1122–1129, Sep. 2003.
217

International Journal of Computer and Electrical Engineering, Vol. 3, No. 2, April, 2011
1793-8163
[14] Y. Jang and M. M. Jovanovic, “A new family of full-bridge ZVS
converters,” IEEE Trans. Power Electron., vol. 19, no. 3, pp. 701–708,
May 2004.
[15] M. Borage, S. Tiwari, and S. Kotaiah, “A passive auxiliary circuit
achieves zero-voltage-switching in full-bridge converter over entire
conversion range,” IEEE Power Electron. Lett., vol. 3, no. 4, pp.141–
143, Dec. 2005.
R. Samuel Rajesh Babu has obtained his B.E
degree from Madras University in 2003. He has
obtained his M.E degree from Anna University in
2005. Presently he is doing his research at
Sathyabama University. His area of interests is
DC - DC converters
Dr. Joseph Henry has obtained his B.E Degree
from Madras University in 1960.He obtained his
M.E degree from IIT-Bombay in 1964. He
obtained his Ph.D degree from IIT Delhi in
1978. Presently he is a professor in Vel Tech
University. His areas of interest are Power
Electronics and Digital Protection
218