In-plane cavity-backed coplanar waveguide to rectangular ...

www.ietdl.orgPublished in IET Microwaves, Antennas & PropagationReceived on 23rd April 2011Revised on 26th July 2011doi: 10.1049/iet-map.2011.0194In-plane cavity-backed coplanar waveguide torectangular waveguide transitionM. Vahidpour K. SarabandiISSN 1751-8725Radiation Laboratory, Department of Electrical Engineering and Computer Science, University of Michigan, Ann Arbor,Michigan, USAE-mail: mvahid@umich.edu; saraband@eecs.umich.eduAbstract: A transition from cavity-backed coplanar waveguide (CBCPW) to rectangular waveguide using a short-circuited pin inconjunction with resonant sections of coplanar waveguide (CPW) and reduced height waveguide segments is presented. Theproposed transition is designed with prismatic features in order to ease the fabrication process. An equivalent circuit model isdeveloped to facilitate the design procedure and full-wave analysis is used to optimise the design. As an example a transitionoperating at Ka-band is designed and shown to have insertion loss of less than 1 dB and reflection below 210 dB over 9% ofbandwidth for a back-to-back structure. To further validate the results a prototype transition at Ka-band for a standard WR-28rectangular waveguide was fabricated and tested. The measured S-parameters of the prototype show a very good agreementwith the simulation results.The increasing demand for microwave and millimetre-wavemonolithic integrated circuits has led to the developmentand implementation of coplanar waveguides (CPWs). CPWlines are commonly used at millimetre-wave frequenciesbecause of their compact planar geometry and ease ofintegration with shunt and series elements which eliminatesthe need to process backside and the use of via holes. Inaddition, the design of microwave probes for on-wafercharacterisation of millimetre-wave-integrated circuits isanother important motivation to employ CPW lines. However,in all planar transmission lines, loss increases by frequency.On the other hand, rectangular waveguides are widely used athigher frequencies as antenna feeds and filters because of theirhigh Q-factor, low-loss characteristics and high-powerhandling capabilities. Owing to these features, it is common tosee submillimetre- and millimetre-wave devices combinewaveguide components with CPW lines. For such devices,obviously an appropriate low-loss transition is required.Various microstrip-to-waveguide [1–3] and CPW-torectangularwaveguide transitions have been proposed in thepast. These transitions are either along or perpendicular tothe direction of propagation. The cosine tapered ridge [4, 5]and ridge-through waveguide [6] transitions along thepropagation direction of the waveguide have been studied atX- and Ka-bands and shown to be broadband while theysuffer from their electrically large dimensions. Another setof broadband designs are proposed based on a transitionfrom a rectangular waveguide to antipodal finline [7], whichalso employ an electrically long substrate in the waveguide.This transition also requires an additional finline to CPWtransition. In [8], by employing a tapered slotline probe anda quarter-wavelength impedance transformer, a unifiedtransition is proposed and demonstrated at X-band. Thistransition could be realised irrespective of the permittivityof the substrate, but has a limited bandwidth for the highpermittivitysubstrates [9]. To overcome this problem, in[10], a multi-section transformer is employed for widebandmatching. Recently, a uniplanar transition based on thetapered finline has been reported with an insertion loss ofless than 1 dB with 9% bandwidth at Ka-band [11]. Thesetechniques, however, involve a high degree of fabricationcomplexity which cannot be easily adopted to silicon orother types of microfabrication techniques.The transitions transverse to the direction of propagationare mainly based on the extension of a suspendedresonant probe from one guiding structure to another.CPW-to-waveguide transitions are similar to coaxial line-towaveguidetransitions [12–14] while the dimensions of thesuspended probe and the distance to the backshort have to becarefully designed and optimised. The common feature of allthese transitions is suspended probes or other 3D structuresinside the waveguide which are not amenable formicrofabrication. A transition using uniplanar quasi-Yagiantenna has been reported for wide bandwidth and easyintegration at X-band, but suffers from utilising a highpermittivitysubstrate in order to fit the antenna inside therectangular waveguide [15]. In the end-launcher aperturecoupledapproach [16], the CPW line with the substrate iscentred in the waveguide with the centre conductor formedan L-shaped loop to couple power to the waveguide mode.As the frequency increases and the dimensions of the lineand waveguide shrink, conventional machining techniquesfail to provide the required tolerance. The literatureconcerning mirofabricating waveguide structures at W-bandand higher frequencies is rather sparse. There have beenseveral attempts to fabricate W-band waveguides withIET Microw. Antennas Propag., 2012, Vol. 6, Iss. 4, pp. 443–449 443doi: 10.1049/iet-map.2011.0194 & The Institution of Engineering and Technology 2012

www.ietdl.orglow-cost microfabrication techniques such as lithography [17,18]. However, in these techniques, the height of thewaveguide is limited by the maximum thickness of the spunphotoresist, limiting the fabrication to the reduced-heightwaveguides which suffer from high attenuation. Takingadvantage of the ‘snap-together’ technique, a rectangularwaveguide was fabricated in two halves and then the halveswere put together to form a complete waveguide [19–21].An alternate technique to etch the waveguide is deepreactive ion etching (DRIE) of silicon. Unlike wet etchingwhich is dependent on the crystal planes of silicon, DRIE isisotropic and provides vertical sidewalls. Hence, DRIE is aviable approach for fabrication of a high-performancemicromachined waveguide structure. In [22, 23], a transitionusing microfabrication processes with separately fabricatedand assembled probe has been reported for both diamondand rectangular waveguide with 20% bandwidth. Anotherhigh-precision silicon micromachined transition with acapability to integrate filters has been proposed in [24, 25]and shows wideband characteristics at the same frequencyrange. However, limitations of microfabrication processesdo not allow fabricating many of the aforementionedtransitions because of the complexity of the geometriesand the number of steps needed in their assembly.In this study, we propose an in-plane cavity-backedcoplanar waveguide (CBCPW) line-to-rectangular waveguidetransition with prismatic features that does not requiremultiple parts and complex assembly. In this approach, theneed for fabricating suspended probe is eliminated andeffective transition is achieved using two resonant structures,namely, shorted CPW line over the waveguide followed by anE-plane step discontinuity. Since the design is very simplewith the features aligned with the Cartesian coordinate planes,it is highly compatible with microfabrication processes.However, it should be mentioned that since the design isusing short-circuited pin, the transition bandwidth issomewhat limited. The transition is modelled by an equivalentcircuit to help with the initial design which is then optimisedusing a full-wave analysis. To demonstrate the validity of thetransition and its model, a back-to-back structure is fabricatedby conventional machining methods at Ka-band and themeasurement results are compared to the simulations.1 Design considerations of waveguidetransitionsTraditional transitions based on E-plane probe excitation ofthe waveguide mode involve attaching a suspended probeto the centre conductor of CPW or coaxial linesperpendicular to the broad wall of waveguides as shown inFig. 1a. The suspended probes in waveguide can be viewedas an infinite array of dipoles and thus can be matched rathereasily to a coaxial line and provide octave bandwidth.The coaxial line-to-waveguide transitions are commonlyused at microwave frequencies since fabrication andassembly is rather straightforward. At high millimetre-waveand submillimetre-wave frequencies where dimensions arevery small, suspending small metalised probes and assemblywith the required tolerance is a challenging task. If the probeis extended, all the way to the lower plate of the waveguideto form a shorting pin then the pin can be formed on samesubstrate and can be bonded to a top substrate to make therequired electric connection. However, a short-circuitedprobe is not resonant, acts purely reactive and cannot bematched to the CPW line directly. In the next section, thearchitecture needed for making the transition from CPW torectangular waveguide using a shorting pin is described.2 Transition using a shorting pinTo properly excite a waveguide with a shorting pin, a resonantcondition must be achieved to eliminate the reactance of thepin. It is well known that a pin terminated by the broadwall of a rectangular waveguide acts as an inductiveelement whose inductance is inversely proportional to itsdiameter and the waveguide dimensions [26]. The geometryand the equivalent circuit model of a shorting pin areshown in Fig. 1b. For this case the transformer turn ratiocan be calculated from [27]√( )2a tan ka 2n =b ka(1)Fig. 1Design considerations of waveguide transitionsa Traditional resonant probe excitation for waveguideb Probe is terminated by the broad wall of the waveguide. In the equivalent circuit mode, L p is the equivalent inductance (X p ), C b is the series capacitance of theshort-circuited probe (X b ) [26] and Z 0 is the characteristic impedance of the waveguidec E-plane step discontinuity generates capacitances required for resonance needed for the mode conversion c the equivalent circuit model for the proposedresonanced The equivalent circuit model for the proposed resonance, X step is the equivalent capacitance of the step discontinuity444 IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 4, pp. 443–449& The Institution of Engineering and Technology 2012doi: 10.1049/iet-map.2011.0194

www.ietdl.orgwhere a and b are the width and height of the rectangularwaveguide, respectively, and k is the free spacewavenumber as described in [27]. The reactance X can beestimated by variational methods [27] and is capacitivebelow and inductive above the resonant frequency. Tocompensate for the inductance of the shorting pin X p , acapacitive element is needed. Since a step discontinuity inthe E-plane of the waveguide acts as a capacitive element, itcan be used to compensate for the inductive behaviour ofthe pin. That is, a resonant condition can be realised byterminating a short-circuited pin in a reduced-heightwaveguide with a step transition from the reduced-heightwaveguide to the standard-size waveguide as shown inFigs. 1c and d. The length of the waveguide between thepin and the step transition can be used to control thecapacitance seen by the inductance. Also, the waveguideheight can be used to control the capacitance at the steptransition point. This structure eliminates the use ofsuspended probe for transition and is simple to fabricate.3 CBCPW to rectangular waveguidetransitionCBCPW lines are preferred at very high frequencies formounting active components because of their low-losscharacteristics [28] (Fig. 2). Hence, a transition from alow-loss membrane-supported CBCPW to rectangularwaveguide is considered here. In this CBCPW, since thedielectric substrate is removed and the line is suspendedover a hollow trench, dielectric loss has been eliminated.Also, the conductor loss is decreased substantially byreducing the current density near the edges and distributingit more uniformly over the metallic strip and the groundaround it. For fabrication purposes, a dielectric membraneon top of the line supports the suspended line over thetrench. This line can be easily incorporated with hollowrectangular waveguides [28].To design the transition based on the idea of the shortcircuitedpin, first the dimensions of waveguide andCBCPW line are chosen based on the desired frequencyrange. The initial values of elements of the circuit modelare selected using the analytical formulas and measurementresults reported in [26, 27]. These values along with thelength of waveguide and CPW line sections are optimisedusing transmission line analysis of the circuit model toobtain the resonant behaviour. A structure based on thesevalues is designed and then optimised using full-wavesimulator Ansoft HFSS.The proposed transition from CBCPW to rectangularwaveguide is presented in Figs. 3a and b. In this transition,three conversion steps are required: from CBCPW toconventional CPW inside the waveguide, CPW to reducedheightwaveguide and reduced-height waveguide tostandard waveguide. The CPW line is narrowed as it entersthe reduced-height waveguide in order to create atransmission line resonator that includes the pin. A secondresonator is created between the pin and the stepdiscontinuity inside the waveguide. The centre conductor ofthe CPW line is open circuited at the location of the pinand the pin is connected to the lower wall of a reducedheightwaveguide. On the other side of the pin, thereduced-height waveguide is short circuited at a distance toappear as another reactance parallel to the pin inductance asshown in the equivalent circuit model in Fig. 3c. Thediscontinuity from CBCPW line to CPW line inside thewaveguide is also modelled by a shunt capacitance.To demonstrate the performance of the proposed transitiona Ka-band prototype is considered. The equivalent circuitmodel parameters are extracted as outlined before andthe S-parameters of a back-to-back structure from CBCPWto rectangular waveguide and rectangular waveguide toCBCPW based on the circuit model are shown in Fig. 3d.This design offers almost 9% of the bandwidth and lessthan 1 dB insertion loss. At the next step, Ansoft HFSS isused to analyse and optimise these values by adjustingthe lengths of transition line segments, the step height andthe pin thickness. The reflection coefficient of the optimisedstructure is compared to that of the circuit model for theback-to-back transition in Fig. 4. As shown, a goodagreement is obtained. It is also shown that the physicalstructure provides a better performance than the onepredicted by the circuit model. The electric field distributionof the optimised structure is shown in Fig. 5a andwell represents the conversion of the CPW mode tothe waveguide mode using the terminated pin. Fig. 5brepresents the reflection and transmission coefficients of thistransition. It is observed that a transmission above 21 dBand reflection below 210 dB for more than 9% ofthe bandwidth can be achieved. An enlarged plot of thetransmission coefficient over the passband is also presentedin Fig. 5c.3.1 Bandwidth enhancementThe bandwidth of the transition is sufficient for mostapplication at these very high frequencies. However, insituations where more bandwidth is needed, the number ofshorting pins and step discontinuities can be increasedaccordingly.To add one more resonance, another shorting pin and stepdiscontinuity can be introduced. The geometry of a three-polestructure is shown in Fig. 6a. Since the height of the stepdiscontinuity is smaller which introduces smallercapacitance values, the pin must be thinner to provide aresonance value close to the previous ones. The dimensionsare optimised using Ansoft HFSS and the reflectioncoefficients of the structure are presented in Fig. 6b. Itisshown that the additional resonance increases the bandwidthup to about 15%.4 Fabrication and measurementFig. 2 Low-loss membrane supported cavity-backed CPW [28]In this line, the dielectric substrate is eliminated and the centre conductor issupported by a membrane on top. The transition of this line to rectangularwaveguide is investigatedAs mentioned before, a dielectric membrane is used tosupport the top suspended thin metal layer of CBCPW andprevent it from collapsing. The membrane covers theconductor and the structure cannot be fed by probes fromIET Microw. Antennas Propag., 2012, Vol. 6, Iss. 4, pp. 443–449 445doi: 10.1049/iet-map.2011.0194 & The Institution of Engineering and Technology 2012

www.ietdl.orgFig. 3Proposed transition from CBCPW to rectangular waveguidea CBCPW to rectangular waveguide transition. It includes a transition from CBCPW to CPW, CPW to reduced-height waveguide mode and reduced-heightwaveguide mode to standard waveguide modeb Top view, side view and a side view of the back-to-back structurec Equivalent circuit model for the transition for Z CBCPW ¼ 50 V, Z CPW ¼ 85 V, X t ¼ 2j 140 V, X ¼ 2j 110 V, n ¼ 1, X b ¼ 2j 8 V, X p ¼ +j 58 V,X step ¼ 2j 1000 V, l backshort ¼ 5mm¼ l g /3, l 2 ¼ 10 mm ¼ 2l g /3, l 3 ¼ 17 mm. Waveguide dimensions are set as the standard dimensions of WR-28Ka-band waveguidesd Reflection and transmission coefficients of the circuit model of the back-to-back structure for the initial values of the elements to achieve a low conversion lossFig. 4 Optimised transition with HFSS is compared to the initialdesign based on the circuit model for a back-to-back transitionand shows good agreement and improved performancethe top. Basically the top layer and the membrane areprotecting the active components that excite or receive thewaveguide signal. In order to test the transition and be ableto make contact to the probe, the top layer of the membraneis also patterned with a CPW line and is connected to thebottom layer through via holes as shown in Fig. 7. Withthis transition, the structure can be fed from the top by50 V GSG probes.To fabricate this structure at Ka-band, the lower partsincluding the CBCPW, reduced-height waveguide andstandard-height waveguide are machined on an ultramachinablealloy 360 brass plate and are gold-plated toprevent oxidation. Rogers Duroid 5880 with 1/4 oz (8mm)electrodeposited copper foil is used as the top cover of thewaveguide as well as the membrane for CBCPW. Thethickness of the substrate is chosen to be 0.127 mm (5 mil),thin enough to act as the membrane. Both sides of thesubstrate are patterned and connected by metalised via holesas shown in Fig. 7. Additional vias are drilled in thesubstrate in order to suppress the substrate mode. Thefabricated parts are represented in Fig. 8.446 IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 4, pp. 443–449& The Institution of Engineering and Technology 2012doi: 10.1049/iet-map.2011.0194

www.ietdl.orgFig. 5The optimised structurea Electric field distribution at 30 GHz for a back-to-back transition with S 1 ¼ 1.46 mm, W 1 ¼ 0.37 mm, S 2 ¼ 0.972 mm, W 2 ¼ 0.247 mm, W WG ¼ 7.112 mm,h WG ¼ 3.556 mm, h 1 ¼ 0.379 mm, h 2 ¼ 1.65 mm, t ¼ 0.49 mm, l 1 ¼ 5.024 mm, l 2 ¼ 9.97 mm and l 3 ¼ 16.8 mmb Side view: E-plane cross section (waveguide centre at maximum E-field)c Reflection and transmission coefficients for the back-to-back transition at Ka-bandd Transmission coefficient at the passbandDifferent techniques were tested to bond the substrate to thebrass unit. Initially solder paste and silver epoxy were used.However, since a high temperature is needed for pastesand epoxies, the substrate was warped and damaged, andthe brass unit was oxidised. Also, the results were notsatisfactory because of misalignments and air gaps aroundthe edges. In addition, the solder reflow caused shortcircuits in the substrate during the bonding process.Moreover, many times the thin pins were bent and easilyFig. 6Bandwidth enhancementa Proposed three-pole structure for enhanced bandwidthbS-parameters of the transitionBandwidth of the transition is increased to about 15% as a result of having athree-pole structure by adding one resonance, a shorting pin and a stepdiscontinuity with h 3 ¼ 2.26 mm, l 1 ¼ 5 mm, l 2 ¼ 10 mm, l 4 ¼ 3.3 mm,l 5 ¼ 6 mm. Additional resonances can be introduced for further enhancement.broken during bonding at high temperatures. The best resultswere obtained by screwing the two units together. Since thesubstrate is very thin, a thick metal plate is used on top of itto ensure that sufficient pressure is applied to the substrateand brass unit to prevent possible air gaps. The silver-platedlong rigid steel wires are snapped into holes drilled on thebottom of the reduced-height waveguide and soldered. Thepins pass through holes cut on the substrate to facilitatealignment of the substrate and the unit.After bonding, the structure is fed by ground-signal-ground(GSG) probes connected to the network analyser using Kabandflexible cables. Prior to measurement it is essentialthat the probes and cables be calibrated. For this purpose,on-substrate thru-reflect-line (TRL) calibration lines aresimulated and designed. For TRL calibration, threecalibration standards are fabricated on the same substrate.These include two line segments with known phase shiftsfor thru measurements and an open-ended line with aknown electrical length for reflection measurement. Aftercalibration, S-parameters of the transition are measured andpresented in Fig. 9. The measurement results show a goodagreement with the simulation, around 9% of the bandwidthbelow 210 dB for return loss and over 21 dB for insertionloss. The observed deviations from the simulation areattributed to misalignment, especially near the pin positionsand possible air gaps between the substrate and the brassunit. The air gap should be of lesser concern since screwsconnecting the substrate to the brass unit should applyenough pressure to make good metal-to-metal contact.Since alignment is done with hand, small misalignment canlead to small deviations from the ideal response. Toexamine this hypothesis, a simulation was carried out wherethe top substrate was shifted to one side by a very smallamount to produce about 300 mm translational displacementat the pin position. This is a typical length that might bemisaligned by manually aligning. Fig. 10 shows theIET Microw. Antennas Propag., 2012, Vol. 6, Iss. 4, pp. 443–449 447doi: 10.1049/iet-map.2011.0194 & The Institution of Engineering and Technology 2012

www.ietdl.orgFig. 7CPW to CBCPW transitiona Transition from a 50 V CPW line patterned on top of the substrate to the CBCPW line patterned on the bottom layer. Lines are connected through electroplated via holesb Side viewFig. 8Fabricated partsa CBCPW trench, reduced-height waveguide and the standard waveguide are machined on a brass plateb Top cover includes the lines is a substrate whose both sides are etched with CPW patternsFig. 9Simulation compared to measurement resultsa Reflection coefficientb Transmission coefficientBoth show a good agreement with the simulation. Observed deviations from the simulation are attributed to misalignment, especially near the pin positions andpossible air gaps between the substrate and the brass unitsimulation results for the misaligned structure where it isshown that the transmission has decreased specifically forfrequencies lower than 29 GHz. Also, the impedance matchnear 31 GHz is not as good as what the structure is capableof producing. This is very similar to the measured resultsindicating that even careful hand alignment is not goodenough to achieve the best performance this transition canproduce at Ka-band. To achieve better results, a betteralignment approach will be needed. For the ultimateapplication of this transition at Y-band wafer alignmentapproaches in clean rooms are available for achievingalignment accuracy down to 1 mm.448 IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 4, pp. 443–449& The Institution of Engineering and Technology 2012doi: 10.1049/iet-map.2011.0194

www.ietdl.orgFig. 10 Simulation results of a structure with misalignments (300 mm translational displacement) between the top substrate and the brass unitcompared to the measured resultsTransmission is decreased for frequencies lower than 29 GHz and the impedance match near 31 GHz is not as good as that of the optimised5 ConclusionWe demonstrated a very simple and easy-to-fabricateCBCPW line to rectangular waveguide transition withprismatic features. The transition is based on the idea ofresonance excitation but does not include a suspendedprobe which was involved in the previous transitions.Using a short-circuited pin makes the transition amenablefor machining and micromachining techniques with finetolerance needed for millimetre- and submillimetre-waveapplications. An equivalent circuit model, in which lumpedelements model the discontinuities and transitions, wasdeveloped for the initial design. A back-to-back structurefrom CBCPW to waveguide and waveguide to CBCPWwas then designed, optimised and fabricated at Ka-band tovalidate the results. The insertion loss below 1 dB andreflection of less than 210 dB over more than 9% ofbandwidth has been achieved. The bandwidth enhancementwas shown by introducing an additional resonance.6 References1 Reig, C., Navarro, E.A., Such, V.: ‘Full-wave FDTD design and analysisof wideband microstrip-to-waveguide transitions’, Microw. Opt.Technol. Lett., 2003, 38, pp. 317–3202 Grabherr, W., Huder, W.G.B., Menzel, W.: ‘Microstrip to waveguidetransition compatible with MM-wave integrated circuits’, IEEE Trans.Microw. Theory Tech., 1994, 42, pp. 1842–18433 Kim, J.P., Park, W.S.: ‘Network analysis of inclined microstrip-slotlinetransitions’, IEE Proc. Microw. Antennas Propag., 2000, 147,pp. 412–4164 Ponchak, G.E., Simons, R.: ‘A new rectangular waveguide to CPWtransition’. IEEE Microwave Theory Tech. Int. 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