ZTE Communications

S pecial Topic
Greater than 200 Gb/s Transmission Using Direct-Detection Optical OFDM Superchannel
Wei-Ren Peng, Itsuro Morita, Hidenori Takahashi, and Takehiro Tsuritani
10 GHz passbands targeting one of the carriers and the
desired signal band. Because of the power loss of DPF, an
EDFA and an 80 GHz optical bandpass filter (OBPF) raise the
signal power before the signal enters the photodiode. In this
experiment, the first to fifth bands are demodulated with the
left carrier (carrier 1), and the sixth to ninth bands are
demodulated with the right carrier (carrier 2). After the
photodiode, the desired band is down-converted to its
baseband via an electrical I/Q demodulator. This I/Q
demodulator comprises one splitter, one synthesizer, one
power amplifier, two mixers, and two electrical low-pass
filters (3 dB bandwidth = 3.7 GHz). The I/Q output signals are
recorded by a real-time scope operated at 20 GSa/s.
Synchronization, cyclic prefix removal, channel estimation,
and equalization are performed offline using MATLAB. The
BER is evaluated with an error-counting method, and for each
BER analysis, 2 million sampling points are considered.
Fig. 3(a) shows the optical spectra of the transmitter output
(resolution = 20 MHz); Fig. 3(b) shows the DPF output
(targeting the 5th band); and Fig. 3(c) shows the digital
spectrum of the 5th band after the real-time scope (where the
band index is defined in Fig. 3a). In Fig. 3(c), the power
ripples on top of the signal come from the power reflection, at
the RF ports, of the mixers in the I/Q demodulator. These
ripples are a function of frequency and introduce some OSNR
penalty to the system, that is, implementation penalty.
Here, we highlight several points about the experiment
setup. First, at the transmitter, the length of the optical paths
between the sideband and carrier branches should be
controlled so that they are as similar as possible. A significant
difference in length would lead to strong phase incoherency
between the carrier and sideband, and this would cause
dramatic phase noise after the photodiode. In this experiment,
we not only equalized the optical lengths of the carrier and
sideband paths, but we also used a zero-overhead phase
noise compensator at the receiver (section 5). Second,
6.5 GHz and 19.5 GHz frequencies for the 9-comb generator
are phase-locked in order to maintain the orthogonality and
reduce linear crosstalk between the adjacent bands [13].
However, because of the sufficiently large band spacing, the
performance is hardly degraded, even if we remove the phase
locking between the synthesizers. Third, to obtain a
broadband bandwidth, we assemble the I/Q demodulator with
discrete components rather than use an integrated I/Q mixer,
which would introduce I/Q imbalances into the signal. This
would mean I/Q imbalance estimation and compensation
would be critically necessary in our system. In section 5, we
introduce a low-overhead I/Q imbalance estimation approach
that uses only two training symbols, that is, the same symbols
for channel estimation.
5 Low-Overhead Signal Processing
Methods
To compensate for I/Q imbalance and phase noise, we
propose a low-overhead training method to estimate I/Q
imbalance and introduce a zero-overhead decision-directed
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ZTE COMMUNICATIONS
March 2012 Vol.10 No.1
phase noise compensator (DD-PNC) [19].
5.1 I/Q Imbalance Training Method
For each frame, the proposed method uses only two
consecutive training symbols (the same as those for channel
estimation). The first symbol is randomly generated, and the
second symbol simply copies the first one and inverts the
signs of the data symbols on negative subcarriers. The two
consecutive training symbols are denoted [ak, b-k] and
[ak, -b-k], where ak and b-k are the data symbols on k th
and -k th subcarriers, respectively, and k is a positive integer
ranging from 1 to Nd/2, with Nd being the data subcarrier
number. At the receiver, the received training symbols
disrupted by I/Q imbalances are denoted [pk, q-k] and [rk, l-k]
for the first and second training symbols, respectively. Then,
the input and output symbols can be expressed in 2 × 2
mutually coupled matrixes [20]:
pk
q-k
and
rk
l-k
H 11 H 12 ak
= (4)
H 21 H 22 b-k
H 11 H 12 ak
= (5)
H 21 H 22 -b-k
The four elements in the 2 × 2 channel matrix H (Hij, which
should contain both the channel response and I/Q
imbalances) can be easily derived with H11 = (pk + rk)/(2a k),
H12 = (pk-rk)/(2bk); H 21 = (q-k + l-k)/(2a k ); and
H 22 = (q-k-l-k)/(2bk), for which the estimation accuracy can be
further improved using the intrachannel frequency-domain
average method [15].
After obtaining the four elements, the inverse matrix of H
can be derived for subsequent equalization. Because this
technique uses the same training symbols (two per frame) for
both channel and I/Q imbalance estimation, the training
overhead is relatively lower than that in [20].
This data-aided adaptive approach is simple, has low
overhead, and is a better way to estimate and compensate for
I/Q imbalance. In the offline I/Q estimation proposal in [21],
the I/Q imbalance in each individual receiver needs to be
customized; however, using this new approach, the I/Q
imbalances of each individual receiver can be handled
adaptively using only two training symbols (which can the
same ones for channel estimation). In particular, the adaptive
method is suitable for lab experiments where I/Q imbalances
vary over time because of the replacement of any component
or the slow bias drift of the optical modulators.
Here we briefly discuss the computational complexity of this
I/Q imbalance estimation method. At the training stage, to
obtain the channel matrix for all subcarriers, 2Nd complex
multiplications are needed for each OFDM symbol, and the
matrix inversions for equalization need an additional 3Nd
multiplications. Therefore, at the training stage, 5Nd
multiplications are needed, which is 4Nd more multiplications
than in regular channel estimation. At the equalization stage,
two subcarriers are jointly equalized via a 2 × 2 matrix so that
the number of multiplications for each OFD symbol is 2Nd,